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WO1996014590A1 - Procede de generation d'une rampe de frequence pour mesurer la duree de propagation de signaux h.f. - Google Patents

Procede de generation d'une rampe de frequence pour mesurer la duree de propagation de signaux h.f. Download PDF

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Publication number
WO1996014590A1
WO1996014590A1 PCT/IB1995/000981 IB9500981W WO9614590A1 WO 1996014590 A1 WO1996014590 A1 WO 1996014590A1 IB 9500981 W IB9500981 W IB 9500981W WO 9614590 A1 WO9614590 A1 WO 9614590A1
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WO
WIPO (PCT)
Prior art keywords
frequency
signal
interrogation
ramp
circuit
Prior art date
Application number
PCT/IB1995/000981
Other languages
German (de)
English (en)
Inventor
Bo A. Fredrikson
Roland Küng
Original Assignee
Tagix Ag
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Tagix Ag filed Critical Tagix Ag
Publication of WO1996014590A1 publication Critical patent/WO1996014590A1/fr

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Classifications

    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03C—MODULATION
    • H03C3/00—Angle modulation
    • H03C3/02—Details
    • H03C3/09—Modifications of modulator for regulating the mean frequency
    • H03C3/0908—Modifications of modulator for regulating the mean frequency using a phase locked loop
    • H03C3/0966—Modifications of modulator for regulating the mean frequency using a phase locked loop modulating the reference clock
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S13/00—Systems using the reflection or reradiation of radio waves, e.g. radar systems; Analogous systems using reflection or reradiation of waves whose nature or wavelength is irrelevant or unspecified
    • G01S13/02—Systems using reflection of radio waves, e.g. primary radar systems; Analogous systems
    • G01S13/06—Systems determining position data of a target
    • G01S13/08—Systems for measuring distance only
    • G01S13/32—Systems for measuring distance only using transmission of continuous waves, whether amplitude-, frequency-, or phase-modulated, or unmodulated
    • G01S13/34—Systems for measuring distance only using transmission of continuous waves, whether amplitude-, frequency-, or phase-modulated, or unmodulated using transmission of continuous, frequency-modulated waves while heterodyning the received signal, or a signal derived therefrom, with a locally-generated signal related to the contemporaneously transmitted signal
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S7/00—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00
    • G01S7/02—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00 of systems according to group G01S13/00
    • G01S7/28—Details of pulse systems
    • G01S7/282—Transmitters
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S7/00—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00
    • G01S7/02—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00 of systems according to group G01S13/00
    • G01S7/35—Details of non-pulse systems
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S13/00—Systems using the reflection or reradiation of radio waves, e.g. radar systems; Analogous systems using reflection or reradiation of waves whose nature or wavelength is irrelevant or unspecified
    • G01S13/74—Systems using reradiation of radio waves, e.g. secondary radar systems; Analogous systems
    • G01S13/75—Systems using reradiation of radio waves, e.g. secondary radar systems; Analogous systems using transponders powered from received waves, e.g. using passive transponders, or using passive reflectors
    • G01S13/751—Systems using reradiation of radio waves, e.g. secondary radar systems; Analogous systems using transponders powered from received waves, e.g. using passive transponders, or using passive reflectors wherein the responder or reflector radiates a coded signal
    • G01S13/755—Systems using reradiation of radio waves, e.g. secondary radar systems; Analogous systems using transponders powered from received waves, e.g. using passive transponders, or using passive reflectors wherein the responder or reflector radiates a coded signal using delay lines, e.g. acoustic delay lines
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S7/00—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00
    • G01S7/02—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00 of systems according to group G01S13/00
    • G01S7/28—Details of pulse systems
    • G01S7/285—Receivers
    • G01S7/288—Coherent receivers
    • G01S7/2886—Coherent receivers using I/Q processing
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03C—MODULATION
    • H03C2200/00—Indexing scheme relating to details of modulators or modulation methods covered by H03C
    • H03C2200/0037—Functional aspects of modulators
    • H03C2200/0058—Quadrature arrangements

Definitions

  • inverters To measure runtimes on SAW elements are inverters. known in the time domain and in the frequency domain.
  • the methods of the second type which are of interest here use a frequency ramp, i. H. a periodic signal with high
  • BEST ⁇ GUNGSKOP-F Frequency change per time interval (sweep). Mixing the transmit signal with the delayed echo signal results in a low-frequency signal with a frequency proportional to the transit time, depending on the transit time.
  • the transmission signal has a constant and - due to the continuous transmission during a certain measuring time - a high transmission energy. The bandwidth is limited by the start and end frequency of the ramp.
  • the coding is e.g. B. in defined phase or Zeit ⁇ shifts or in amplitude modulations.
  • z. B. an interrogation signal with a periodically repeating frequency ramp is used. With such an interrogation signal, the superimposition of the response signal components leads to differential oscillations which are detected by the interrogation station.
  • a system of this type is e.g. B. from US 4,737,790 or US 4,625,207.
  • the object of the invention is to provide a method and a circuit arrangement for the precise, inexpensive and age-free generation of frequency ramps in the RF range (namely in the GHz range).
  • the invention is intended to be particularly suitable for use in the identification of delay-time and / or phase-coded SAW elements.
  • a method according to the invention for generating a frequency ramp for a transit time measurement of RF signals has the following steps:
  • a direct baseband signal is generated with a frequency that increases linearly at least in parts by direct digital synthesis; b) the baseband signal is mixed in a mixer with a carrier signal of constant RF frequency and filtered to extract the frequency ramp (single sideband filter).
  • step b) the baseband signal is given as a reference frequency in a phase comparator of a PLL circuit which generates the RF signal.
  • step a) in a digital memory, respectively.
  • Register temporarily stores an instantaneous value of the frequency for a predetermined step interval and increases or periodically. degraded.
  • the step size is kept so small that (within the scope of the application-specific circumstances) the best possible approximation of the theoretically continuous or constant ramp slope (ie the step-like frequency curve is so fine that it essentially does not affect the running time measurement).
  • phase accumulator In a phase accumulator, a phase of the baseband signal is generated or accumulated at a predetermined clock rate in accordance with the instantaneous value of the frequency.
  • the phase accumulator allows a frequency jump to be carried out without a phase jump. That is, The phase can also be tracked without problems even over sudden frequency changes or signal interruptions.
  • phase is subsequently used as an address for a memory which contains the amplitude values of the corresponding smus and cosine function.
  • stored digital amplitude values are then converted in a D A converter to the analog signal value of the frequency ramp.
  • a sign (cos) or sign (s ⁇ n) amplitude value (sign bit) is stored in a memory corresponding to the generated phase.
  • the square-wave signal generated in this way can be fed as a baseband signal into the mixer or. be given the phase discriminator. In principle, therefore, only the sign of the sine or cosine function is generated and not the exact course of the oscillation. A complex D / A converter is therefore unnecessary.
  • the circuit according to the invention can be implemented completely digitally.
  • a control by very high clock rates makes it possible to interpolate the inaccuracy of the instantaneous frequency caused by the sign function (oversampling).
  • An appropriately designed integrated circuit is preferably used to generate the baseband signal.
  • a function does not require a complex digital / analog part and can therefore be operated at very high clock rates to achieve large bandwidths.
  • An interrogation station which can be used in such an identification system is characterized, for. B. from a transmitter circuit that generates an interrogation signal with at least two, offset by a frequency jump, but phase coherent or with a fixed phase jump to each other ⁇ subsequent linear frequency ramps.
  • the phase adjusting transition allows measurement or Beob ⁇ respectful intervals without enlarging the bandwidth lengthen to ver ⁇ or frequency bands with strong Storern aus ⁇ targeted zublenden.
  • Another possibility for generating different interrogation signals is to provide a control circuit for generating different, in particular pseudo-random, time intervals between the frequency ramps. Interference between query signals and deliberately introduced interference from third parties can be eliminated or minimized in this way.
  • At least two frequency ramps are created immediately after one another.
  • the frequency ramps do not have to be the same length.
  • a longer ramp tui the transponder identification can be a smaller one for setting the AGC (automatic gain control) of the interrogation station be upstream.
  • the two frequency ramps preferably have the same starting requirements and the same slopes.
  • a transmitter circuit is such that an interrogation signal is generated from stucco-linear, phase coherently adjoining frequency ramps, the individual sections having different steepness.
  • an interrogation signal is generated from stucco-linear, phase coherently adjoining frequency ramps, the individual sections having different steepness.
  • two interrogation stations can work simultaneously in the same local and frequency-based transmission range.
  • the interrogation signals belonging to different interrogation stations have different ramp profiles, they can distinguish the RF echo signals reflected by a particular passive transponder. Different ramp steepnesses lead to different differential frequencies. It is also the case that an echo signal whose ramp steepness is not the same of the transmission signal matches, cannot lead to a usable detection signal in the interrogation station after mixing.
  • the choice of the rack parts can thus be used to introduce a type of coding
  • the same result can also be achieved if the start times of a plurality of successive frequency ramps are read device-specific according to a pseudo-random principle, or if pseudo-random transmission pauses take place within a ramp.
  • the phase coherence allows the reader to correctly assemble the signal components if it knows the random sequence itself.
  • a further embodiment of an interrogation station according to the invention is characterized in that a circuit arrangement is provided for detecting a free frequency range and that the starting frequency is placed in the free frequency range found. That is, the identification system does not work with a predetermined signal in a predetermined frequency band, but with flexible variables. This is particularly advantageous if several interrogation stations are arranged in locally partially or completely overlapping transmission areas, which do not necessarily all have to be active at the same time. That is, the RF areas available can be better resp. be used more efficiently. Coexistence with existing narrowband services of other frequency band users is also possible.
  • the object of the invention is to provide a method for generating a linear frequency ramp which can be implemented in a circuit-simple and technically efficient manner.
  • the required precision should be able to be achieved at low cost.
  • the solution is that an integrator of a PLL circuit is optionally supplied with a constant control signal instead of with a phase error.
  • the PLL circuit is initially run as a closed control loop in order to run at a precisely defined starting frequency in accordance with a reference frequency. Only then - to generate the frequency ramp - is the control loop opened and the integrator controlled with the constant control signal.
  • the length of time that elapses between opening the PLL circuit and reaching a stop frequency can be determined (the steepness corresponds to the frequency range passed through per time unit)
  • the constant control signal is increased or decreased in the next cycle. Strictly speaking, the control signal is only constant during a single cycle. However, if the circuit has settled, the control signal will generally only change insignificantly.
  • the PLL circuit is equipped with a switch in front of the integrator in order to open the PLL circuit and to be able to apply the constant error signal to the integrator.
  • An integration constant generator with an adjustable, constant control signal can be electrically connected to the integrator by means of the switch.
  • the circuit arrangement preferably also has a stop frequency detector and a time measurement circuit in order to be able to measure the steepness of the frequency ramp.
  • An automatic correction circuit is preferably provided in order to convert the integration constant generator (constant voltage source, RC element) in accordance with the deviation of the measured ramp steepness to be corrected for the given correction.
  • a manual setting option potentiometer
  • the user can be shown whether the slope is too large or too small
  • a control circuit which controls the switch and a polarity of the integration constant generator so that the output signal of the PLL circuit runs at a constant frequency for a predetermined time, then a first frequency ramp and immediately afterwards a second one Frequency ramp runs through to finally end at the constant frequency mentioned.
  • the signal runs z. B. for a certain time on the start frequency, then increases linearly to the stop frequency and decreases with the opposite steepness to the start frequency.
  • the ascending frequency ramp can be used to detect the code and the descending one to adjust the amplifier (AGC or the like).
  • the increasing and the decreasing frequency ramp do not necessarily have to have the same slope in terms of magnitude. Conversely, the same slopes are circuit-technical and advantageous for the detection of the response signal.
  • a method for detecting a target frequency is used for a signal with a range of increasing or decreasing signal frequency, which is characterized in that the signal is mixed with a predetermined reference frequency in order to generate a differential frequency that by a predetermined small amount df over or; is below the target frequency and that a detection sign is generated as soon as the difference frequency generated is less than the predetermined amount df.
  • the desired target frequency can be easily and technically efficiently detected in terms of circuitry.
  • the method allows the required precision to be achieved at low cost.
  • the reference frequency is not identical to the target frequency, but rather by a precisely defined amount larger (for detecting the target frequency with an increasing frequency ramp) or smaller (im Case of a negative frequency ramp) is selected.
  • the difference frequency between the frequency ramp and the reference frequency will thus initially be very large, but will always become smaller until it has approached the reference frequency by the amount df. At this point in time it is the same size as the specified target requirement.
  • the detection signal therefore shows exactly when the target frequency has been reached or exceeded.
  • the amount df is very small (e.g. 10 - 10 times smaller) than the signal frequency.
  • a counter with a number range of at least N is preferably used.
  • the amount df should correspond to the Nth part of a control requirement f of a control signal.
  • the payer is incremented by the control signal m.
  • the counter is also periodically reset with the difference frequency.
  • the control signal is preferably also used as a reference of a PLL circuit for generating a mixed frequency.
  • the counter is preferably a commercially available, integrated circuit with at least n stages. The number of stages n uss must be at least so large that 2> N. With a counter with n "stages, n"> n, the detection signal can thus be tapped at the nth output.
  • the method is used in particular to control or monitor a signal in the GHz range (e.g. 500 MHz up to a few GHz).
  • the reference frequency is therefore also in this range. If the interrogation signal of a remote identification system moves, for example, in the range from 905 to 925 MHz, the reference frequency is slightly above 925 MHz.
  • the invention is thus used to control or monitor very high signal frequencies.
  • the control frequency is considerably below the signal frequency. Usually it will be below 100 MHz, but will not fall below 100 KHz. A preferred range is between 1 and 20 MHz. Conventional clock signal generators can therefore be used to generate them.
  • df is much smaller than the signal requirement.
  • df will be more than 100 times smaller than the reference frequency.
  • a circuit arrangement with a reference frequency oscillator, a mixer which mixes the signal with the reference frequency for generating the difference frequency, and a detector which generates a detection signal will be required as soon as the difference requirement falls below the defined amount df .
  • the detection of a predefined frequency preferred according to the invention is not limited to the detection of the stop frequency of a frequency ramp. In the case of a linear frequency ramp in the GHz range, as already mentioned, the slope can be determined very precisely by using the invention. (The slope results from the start and end frequency as well as the time required to reach the end frequency.)
  • Another aspect of the invention is a method for contactless interrogation of a passive mobile transponder, wherein an interrogation signal transmitted by an interrogation station is coded back by the transponder as a response signal and is detected by the interrogation station.
  • decoding carried out in the interrogation station and automatic amplifier setting are carried out at separate time intervals. This method is suitable for detecting response signals in the GHz range. It enables the reliable extraction of identification codes from response signals that vary widely in level.
  • a measurement process is thus carried out with a constant measurement gain during a first frequency ramp and the measurement gain is reset (by the AGC) during a second frequency ramp.
  • An interrogation signal is preferably generated, which comprises an ascending and a descending frequency ramp.
  • the interrogation signal is decoded in the course of the one, preferably the ascending, frequency ramp, and the amplifier is adjusted in the course of the other, preferably the, falling frequency ramp.
  • the interrogation signal typically has a cyclical character, ie the rising and falling frequency ramps follow one another alternately.
  • the frequency ramps should be as linear as possible for decoding and for level or amplifier adjustment. However, it is not imperative that the interrogation signal only consist of linear frequency ramps. In particular, e.g. B. after a linearly decreasing frequency ramp, a signal section with a constant frequency is provided. (A transponder that generates the coding with the aid of phase shifts or transit time differences does not lead to a detectable difference signal in the section of constant frequency.)
  • Rising and falling frequency ramps are advantageously of the same size and of the same magnitude in terms of amount (simpler coding or setting of the amplifier). Different curve shapes are, however, not excluded in principle. Differences of 20% with respect to the slope in terms of amount are quite feasible
  • a level detector can continuously process the received and downmixed signal and generate a compensation value which is sampled by a sample-and-hold circuit in the interval reserved for the amplifier setting and temporarily stored for controlling the amplifier for a certain time. In principle, it is a control loop that is periodically spaced (namely during the amplifier setting interval j- is closed. The gain value is kept constant in the decoding intervals, so that no transients can arise.
  • the decoding includes e.g. B. a Fourier transform of the audio signal. This determines code-specific difference frequencies. Keeping the gain value constant in the decoding interval prevents falsification of these difference frequencies by amplitude modulation.
  • the transponder preferably has a SAW unit for coding the signal. In this way, a large number of response signal components with defined relative time delays can be generated from the interrogation signal.
  • a circuit arrangement for carrying out the method has a controllable audio signal amplifier, a downstream decoding circuit and a control circuit which alternately adjusts the amplifier according to the detected level and activates the decoding circuit.
  • the control circuit is driven in accordance with the query signal (synchronous timing).
  • the amplifier setting is therefore always permitted when e.g. B. the linearly descending frequency ramp is generated.
  • An interrogation signal according to the invention for identifying a mobile passive transponder which generates a response signal with time-delayed response signal components, is characterized by a repetitive sequence of frequency ramps that increase and decrease linearly to a comparable extent.
  • a cycle of the signal is preferably divided into a first time range with an increasing and a second time range with a falling frequency. quenzrampe and a subsequent third area with constant frequency.
  • FIG. 1 shows a block diagram of a digital synthesizer according to the invention
  • FIG. 2 shows a block diagram for generating an RF frequency ramp based on the principle of SSB modulation
  • FIG. 3 shows a block diagram of a circuit for generating an RF frequency ramp according to the principle of a PLL circuit
  • FIG. 4 shows a schematic illustration of a system for the contactless scanning of a SAW transponder
  • FIG. 5 shows a schematic illustration of the temporal frequency profile of an interrogation signal with an upstream measuring ramp
  • FIG. 6 shows a schematic representation of the temporal frequency profile of an interrogation signal with the frequency range hidden
  • FIG. 7 shows a schematic illustration of the temporal frequency profile of an interrogation signal with different piecewise linear ramp sections.
  • FIG. 9 is a block diagram representation of a circuit for detecting a response signal of a mobile transponder
  • FIG. 10 shows a block diagram of a circuit arrangement for generating an interrogation signal with a circuit arrangement according to FIG. 9;
  • FIG. 11 shows a schematic illustration of a circuit for generating a linear frequency ramp
  • Fig. 12 is a schematic representation of a circuit for detecting the target frequency.
  • FIG. 1 shows a preferred embodiment of a digital synthesizer 1. It is preferably an integrated module (IC circuit).
  • IC circuit integrated module
  • An external arithmetic and control circuit can load a current frequency value f into a data register 2 in series or in parallel. At a point in time precisely controlled by a command circuit 7, the content of the data register 2 is loaded into a frequency register 3.
  • a phase accumulator 4 increments the phase in accordance with the value contained in the frequency register 3, specifically in the rhythm specified by a clock signal at the clock signal input 8.
  • the phase value generated in this way is output (either directly or via a phase correction element 5) to two ROMs 6a, 6b, which store the signal values cos (2 ⁇ ft) and s ⁇ n (2 ⁇ ft) with a resolution in the amplitude of N bits.
  • the outputs of the two ROMs 6a, 6b are each passed to a D / A converter 9a, 9b in order to generate analog signal values I, Q.
  • the values sign (cos (2 ⁇ ft)) and sign (sin (2nft)) can be output directly as I and Q values at the corresponding outputs. That is, the D / A converter 9a, 9b can be omitted.
  • Data and frequency register 2 respectively. 3 are e.g. B. 32 bit registers that are reloaded for each frequency step of the digital frequency ramp. Alternatively, it is possible to provide several registers in order to store the starting frequency on the one hand and one or more frequency steps on the other hand.
  • a multiplexer connected between the frequency registers and the phase accumulator 4 then forwards the respectively correct frequency value to the phase accumulator 4.
  • the (optional) phase correction element 5 comprises an adder 5a which sums the output of the phase accumulator 4 with a memory content of a register 5b. In this way, the current phase value can be corrected by a constant if necessary.
  • a phase correction can e.g. B. may be necessary to achieve a coherent detection in the echo signal detector.
  • the phase accumulator 4 preferably has a smaller word width at the output than at the input. That is, the 32-bit value read (for example) from the frequency register 3 is, for. B. reduced to a 12 bit value. This is then used to control the two ROMs 6a, 6b.
  • the word width M on the output side of the phase accumulator 4 is of course determined by the desired precision in the zero crossing (maximum phase jitter). For the digital synthesizer 1, a zero crossing can only take place synchronously with the clock signal.
  • An RF oscillator 10 generates in-phase (cos) and quadrature components (sin) of a carrier oscillation of z. B. 2.5 GHz.
  • Two passive diode mixers 11a, 11b which at their local oscillator input with in-phase and quadrature components I and. Q of the digital synthesizer 1 are applied, the corresponding signal components of the carrier vibration modulate. Due to the strongly non-linear characteristic curve of the diodes, the diode mixers 11a, 11b behave practically like an ideal switch mixer.
  • the square-wave signal s ⁇ gn (s ⁇ n) or s ⁇ gn (cos) according to the invention is therefore completely sufficient as a baseband signal for controlling the mixer. (Of course, other baseband signals are also possible.)
  • I and Q components are combined on the output side of the diode mixer 11a, 11b with a sum ierglled 12 and subsequently amplified in the amplifier 13.
  • a bandpass filter 14 filters out the desired frequency range with the frequency ramp for subsequent radiation via an antenna 15.
  • the digital synthesizer delivers a baseband signal in the range of z. B. 40-60 MHz.
  • FIG. 3 shows another possibility of how the RF frequency ramp can be generated with the digital synthesizer 1 already described.
  • the digital synthesizer 1 is in a (relatively low) frequency range from z. B. operated 10-15 MHz.
  • the Q component is compared in a phase discriminator 16 with the signal supplied by a mixer 23 and filtered into a loop filter 17.
  • a VCO 18 is driven with the control signal generated in this way. This generates an output signal in the range of z. B. 2.44-2.46 GHz.
  • the output signal mentioned is fed to an antenna 21 via an amplifier 19 and a bandpass filter 20. But it is also to a prescaler 22, which the frequency z. B. reduced to a quarter, and then mixed in the mixer 23 already mentioned with the signal of a reference oscillator 24 from the intermediate band to the baseband.
  • the reference oscillator 24 operates at 600 MHz.
  • the phase discriminator 16 consists of an EXOR element.
  • the PLL circuit follows the direl-t digitally synthesized frequency ramp precisely and in phase.
  • the signal s ⁇ gn (cos) is applied to the EXOR element, the loop filter 17 takes over the interpolation of the zero crossings, the accuracy of which is determined by the clock signal (jitter).
  • the D / A converter 9a, 9b in the synthesizer 1 according to FIG. 1) can therefore be dispensed with.
  • the frequency ramp is generated by a direct digital synthesis.
  • a digital synthesis with subsequent D A converter is necessary for interpolation of the sample value, or its sign can only be used if the oversampling is correspondingly large.
  • the free programmability of the starting frequency and the ramp steepness permits a versatile application, in particular in contactless interrogation systems with SAW transponders.
  • Fig. 4 illustrates the principle. Under the control of a reading control 26, a reading device 25 generates an interrogation signal which is emitted via an antenna 31.
  • a transponder 32 located nearby receives the interrogation signal with its antenna part 33 and couples it into a SAW element 34. There the signal is implemented code divided into several echo signal components. These are decoupled again from the SAW element 34 and sent back to the reader 25 via the antenna part 33.
  • the signal generated by an interrogation signal generator 28 is not only emitted via hybrid circuit 30 and antenna 31, but is also fed into a mixer 29.
  • the interrogation signal and echo signal are mixed there.
  • a system computer 27 allows the programming of certain interrogation signal variables (for example start frequency, stop frequency, steepness etc.).
  • tag identification systems are explained, which are preferably based on frequency ramps generated according to the invention.
  • tag identification systems mentioned can also be used in a different way, i. H. can be implemented independently of the direct digital synthesis according to the invention.
  • the large frequency ramp Sa. z- rises linearly within an interval I- from a start frequency f n to a stop frequency f.
  • the small frequency ramp S likewise begins at the start frequency f-, but runs, for example, only up to half the stop frequency f / 2.
  • the slopes of the two frequency ramps SS ? are preferably the same size (e.g. 1.25 GHz / s).
  • the interval I. of the small frequency ramp Sa.i is thus half as large as the interval I 2 in the present case.
  • the two frequency ramps adjoin one another, but are separated by a frequency jump. According to the invention z. B.
  • the interval L is the detection interval, ie the echo signals are mixed and analyzed in it with the transmission signal.
  • the setting of the AGC and the interference-sensitive detection can be separated from one another in this way without the bandwidth having to be increased.
  • the interval I- can be set to the same duration as L and the two frequency frames can be connected to one another in a phase-coherent manner. This achieves twice the detection time without the need to increase the bandwidth.
  • a longer interval with a constant frequency can be inserted between the intervals I 2 and I (ie after the detection phase).
  • this interval of constant frequency there is no difference frequency between the transmission signal and the interrogation signal.
  • the frequency ramp starts on an external trigger signal, the receiver input must settle for the sudden level change, with considerable shock response transients occurring.
  • the automatic gain control AGC
  • the system can be provided with an easily metered dose. Accuracy losses due to errors in the starting frequency do not occur. Since the receiver signal processing, the trigger signal and the clock the direct digital synthesis are derived from the same clock signal, the time and frequency of the ramp can be determined or predicted exactly at any time.
  • the observation period for the signal evaluation and transponder identification can easily be extended to several, immediately successive frequency ramps. Due to the exact knowledge of the phase of the transmission signal (the phase is stored in the phase accumulator at all times) at the end of the frequency ramp, it is possible to jump back to the starting frequency in the correct phase and to run through the ramp again. This extends the observation time and thus the temporal resolution of the measurement while the bandwidth of the transmission signal remains the same.
  • the interval for settling is conceptually preceded by the detection interval I 2 .
  • the receiver regulation phase to the detection phase and to freeze the amplification factors between the measurement phases (in particular at intervals of constant frequency and during detection).
  • the interrogation signal S In contrast to the interrogation signal Sa according to FIG. 5, the interrogation signal S, according to FIG. 6, has two frequency ramps S,, and S, - which do not overlap in terms of frequency.
  • the lower frequency ramp S,. is separated from the upper S, - by a frequency band f, f.
  • the frequency hopping takes place in a phase-coherent manner.
  • the frequency band f to f is thus specifically hidden from the echo signal. This represents a possibility in order to be able to selectively hide narrow-band storers or frequency bands that have already been occupied.
  • Wanrend in 6 the intervals, I. directly adjoining one another with the corresponding frequency ramps S, m, S, 2 , time intervals with constant frequency (or without signal amplitude) can also be easily inserted (as in connection with FIG. 5 explained).
  • Frequency ramp S, - (which runs from frequency f to stop frequency f) is equally steep. However, this is not imperative, as the following explanations show.
  • adjusting or changing the ramp slope can also be used to increase the runtime resolution for specific measurements.
  • the steeper the frequency ramp the greater the difference frequency for a given transit time.
  • the observation time specified by a frequency ramp is greater (and therefore the temporal resolution of the measurement is all the better) the lower the slope (as already mentioned, the observation time can be obtained by referring to several frequency ramps can also be extended regardless of the slope of the ramp. )
  • the ramps of the various interrogation stations may be expedient to start the ramps of the various interrogation stations synchronously with one another but precisely with a slight offset (of, for example, 10 KHz).
  • the offset must be greater than the maximum interesting difference frequency (which is, for example, a few kHz).
  • a further variant for differentiating echo signals from different interrogation stations could consist in that the different interrogation stations are operated with suitably differently selected ramp steepnesses.
  • a preferred embodiment is characterized in that the pause intervals of different interrogation stations are of different lengths.
  • the duration of the intervals between successive frequency ramps is e.g. B. determined by pseudo-random codes. That is, Each polling station has its own pseudo random code generator, which defines the length of the intervals mentioned.
  • an interrogation station can in principle also determine a free transmission channel itself and use a frequency occupy the quenzrampe. In this way, a channel-oriented multiple use of a frequency range available can be realized without having to change the hardware. Such a concept is of particular interest if a certain territory can potentially be queried from several locations and if there is always a certain probability that some query stations are inactive.
  • FIG. 8 shows the chronological announcement of the signals relevant for the preferred exemplary embodiment.
  • a stationary interrogation station sends a signal s. which is received by a nearby mobile transponder (tag), is delayed in a defined, code-specific manner and is radiated back. The interrogation station detects the answer signal
  • the interrogation signal s is in accordance with a preferred embodiment. rungsfor the invention periodically Each period is divided into ⁇ divided into three time intervals T .., T. ,, T--.
  • the interrogation signal s has a constant frequency f_.
  • the subsequent second interval T 2 there follows a linear frequency ramp, at the end of which the interrogation signal s. a target frequency f. reached.
  • the subsequent third interval T-. there follows a negative linear frequency ramp, ie the frequency drops from f. on f-. Now the interval T- of the next cycle follows again, etc.
  • the frequency f is z. B. 905 MHz; f is then z. B. at 925 MHz.
  • the rate of increase is typically in the order of 1 GHz / s (eg 1.25 GHz / s).
  • the rising and the falling frequency ramp in the intervals T are the same slope in terms of amount.
  • the interval T is at least as large as the sum of the two intervals T- and T, or is an integer multiple thereof.
  • a preferred working range is also in the range of 2.45 GHz.
  • the response signal s D essentially corresponds to a plurality of differently delayed interrogation signals s ⁇ . This is indicated in FIG. 8 by the dashed lines parallel to the signal s ⁇ .
  • the time delays are e.g. B. in a range of 1 - 10 ⁇ s. That is, , the first component of the response signal appears e.g. B. after about 1.2 ⁇ s, the second after about 1, 3 ⁇ s etc.
  • the code-specific delay is z. B. in the defined deviation from the mentioned 0.1 ⁇ s grid.
  • FIG. 11 shows a ramp generator 120 according to the invention.
  • a reference frequency fref is compared in a phase difference detector 121 with a feedback signal.
  • the circuit according to FIG. 9 works like a conventional PLL circuit.
  • the interrogation signal s runs at a constant frequency.
  • the integrator 124 has a constant control voltage V instead of the phase error signal.
  • V the frequency of the output signal s ⁇ increases linearly.
  • the slope is determined by the size of the control voltage or. by the value of the time constant, which is formed by the RC element of the integrator 124.
  • FIG. 10 shows a schematic illustration of a circuit for generating the interrogation signal s shown in FIG. 8.
  • a reference oscillator 130 supplies the frequency f - f required to control the PLL circuit.
  • the output signal s D is amplified (amplifier 131) on the one hand and emitted via an antenna 102 and on the other hand sent to a stop frequency detector 128. This generates a pulse or an edge as the frequency ramp has reached a predetermined stop frequency.
  • the stop frequency can frequency of the rising frequency ramp or the descending frequency ramp.
  • a control circuit 129 controls the ramp generator 120 as follows:
  • the switch 123 (cf. FIG. 11) is set such that the PLL circuit is closed and the interrogation signal s has a constant frequency f ". After the interval T., the switch 123 is changed over. The interrogation signal s increases linearly, as shown in FIG. 8. As soon as the stop frequency (f.) Is reached, the polarity of the constant control voltage is inverted. The increasing frequency ramp in the interval T is immediately followed by a decreasing frequency ramp T with the same, but negative, slope.
  • the interval T- is the same as the interval T-.
  • the frequency f r is again. of the PLL circuit reached.
  • Switch 123 is switched back to input Ea. switched and the cycle can start again.
  • the control circuit 129 communicates with a superordinate control of the interrogation station in order to match the receiver side to the interrogation signal s n shown in FIG. 8.
  • the special design of the query signal s. is used in an advantageous manner on the receiver side.
  • the response signal s D generated by the transponder is decoded. This is illustrated by the control signal S in FIG. 8.
  • the amplifier setting takes place in the falling frequency ramp, that is to say in the interval T- (for example with an AGC circuit). This is illustrated by the signal S .. in Fig. 8, which goes high only in the interval T-.
  • the aforementioned implementation of the amplifier setting and decoding at two different intervals has the advantage that transients which arise during the amplifier setting are kept away from the decoding interval.
  • the invention is of course not limited to the exemplary embodiments explained.
  • the circuit according to the invention can be used wherever a linear frequency ramp with a precise starting frequency has to be generated with little effort.
  • the integration constant can vary the slope of the frequency ramp. be corrected.
  • the time is measured which elapses before the frequency of the interrogation signal s ⁇ from the starting frequency f 'to the target frequency f. has risen.
  • the measured time is compared with a specified time range. Depending on whether the measured time is too short or too long, the integration constant is increased or decreased accordingly.
  • the error can be displayed to the user (e.g. via LEDs) so that he can adjust the integration constant manually (e.g. by adjusting a potentiometer). If an automatic control is desired, z. B. a PI controller can be used.
  • the accuracy of the start frequency and the measurement accuracy of the rise time depend on the accuracy of the reference oscillator 130 of the PLL circuit. It is possible to limit the frequency fluctuations in the entire temperature range of the oscillator to less than 1.5 ppm. The accuracy of the stop frequency naturally depends on the accuracy of the frequency detector. At signal frequencies in the GHz range (e.g. 905 MHz or 2.45 GHz), the error is typically less than ⁇ 10 - ⁇ 20 KHz. It follows from this that the maximum relative error of the average slope of the frequency ramp can be kept at ⁇ 0.1-0.2% in the entire temperature range.
  • the average signal level detected by the interrogation station fluctuates greatly.
  • the envelope of the signal to be detected is not constant and is like a noise-like signal.
  • the time constant of the detector must be smaller than T-.
  • the mean value also fluctuates.
  • the S + H circuit ensures that a current mean value is frozen and remains constant for the duration T-.
  • FIG. 9 A block diagram of a switching arrangement for detecting the response signal s will now be explained with reference to FIG. 9.
  • the interrogation station is designated by 101. It comprises an interrogation signal generator 104, which generates the interrogation signal s shown in FIG. 8.
  • the interrogation signal s ft is emitted via an antenna 102.
  • a mobile transponder (SAW tag), not shown in detail, reflects a coded response word signal s R. This is received by an antenna 103 and fed to a front-end circuit 106 known per se.
  • a mixer 107 mixes the received signal with the aid of a local oscillator LO into a baseband in the audio signal range.
  • the downmixed signal is filtered in a manner known per se using a bandpass filter 108 and amplified by an amplifier 109.
  • a third amplifier 111 generates the output signal suitable for decoding (decoding circuit 115).
  • a level detector 112 which is connected upstream of an integrator 113, also hangs at the output of the amplifier 111.
  • the mean signal level of the audio signal determined in this way is used to control the amplifier 110.
  • the control circuit 105 also controls the decoding circuit 115 and the interrogation signal generator 104.
  • the signal S for controlling the S + H circuit 114 remains in the intervals T. and T 2 to low and is only in the interval T-. (Set to high, in particular in a time-centered partial area from TM. Accordingly, the stored value, in particular during the interval T ?, is held (H). Only in the interval T is a new value read in (S).
  • the decoding circuit 115 is only activated in the interval T 1.
  • the signal S is therefore only in the interval T ? set to high.
  • the ACCORDANCE OF INVENTION ⁇ the signal decoding and Verstarkereingnagnagnagnagnagnagnagna in two different intervals performed.
  • the invention is of course not limited to the exemplary embodiments described.
  • the circuit arrangement according to FIG. 9 can be expanded or modified in various ways.
  • the query signal shown in FIG. 8 does have two mirror-image frequency ramps, but this is not a mandatory feature.
  • the frequency ramps can be of different steepness or generally have a different course (exponential sinusoidal etc.).
  • both the decoding and the amplification take place within a "frequency sweep". It is important that the two functions are separated in such a way that transients can interfere with the decoding due to the amplifier setting. Furthermore, the amplifier setting should not be separated from the decoding too much in terms of time
  • FIG. 12 shows a block diagram of a preferred embodiment of a stop frequency detector 128.
  • the interrogation signal s A at the output of the ramp generator 120 is passed to a mixer 142 on the input side via a voltage follower 141.
  • a low-pass filter 143 at the output of the mixer 142 filters the sought-after difference frequency f.
  • a subsequent trigger circuit 144 generates a square-wave signal which is given to a RESET input 148 of a payer 145.
  • a clock signal input 149 is supplied with a control signal of the frequency f of a clock signal generator 147.
  • the detection signal 7 is tapped at an output Q corresponding to the nth stages of the counter 145. It changes its state if the target frequency has never been reached
  • the Ab raqesignal s A is mixed with a reference frequency fr, which is generated by a PLL circuit 146 This is controlled by the control signal with the frequency f of the clock signal generator 147.
  • the frequency of the interrogation signal s begins at a start frequency f and rises linearly against the stop or target frequency.
  • the task of the stop frequency detector 128 is to determine when the target frequency f .. has been reached.
  • the reference frequency f is determined as follows:
  • df 312.5 KHz
  • f 1 is z. B. 925 MHz and the start frequency z. B. 905 MHz.
  • the difference frequencies f is thus a maximum of 20.3125 MHz and decreases as the frequency ramp rises to 312.5 KHz.
  • the difference signal and the control signal do not run synchronously, there is an area of uncertainty in practice.
  • n in the range 4-9 is recommended for the signal parameters given as examples.
  • f 12.8 MHz z.
  • B. has (with a ramp c

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  • Engineering & Computer Science (AREA)
  • Radar, Positioning & Navigation (AREA)
  • Remote Sensing (AREA)
  • Computer Networks & Wireless Communication (AREA)
  • Physics & Mathematics (AREA)
  • General Physics & Mathematics (AREA)
  • Radar Systems Or Details Thereof (AREA)

Abstract

Afin de générer une rampe linéaire de fréquence lors de l'identification de répondeurs passifs mobiles d'identification à ondes de surface, on génère un signal de bande de base à fréquence ascendante linéaire par synthèse numérique directe (1). Ce signal est ensuite modulé sur une fréquence porteuse dans le domaine des GHz par des oscillateurs H.F. et des mélangeurs. De préférence, on détermine avec un accumulateur de phase une phase correspondante à une valeur momentanée de fréquence et on mémorise le signe d'une fonction sinusoïdale ou cosinusoïdale sur la base de la phase ainsi déterminée. Le signal rectangulaire ainsi obtenu représente le signal de la bande de base. Une station d'interrogation génère des signaux d'interrogation constitués de plusieurs rampes de fréquence qui ont des phases cohérentes, qui se suivent à des sauts de phase prédéterminés ou qui sont séparés par des sauts de fréquence.
PCT/IB1995/000981 1994-11-08 1995-11-08 Procede de generation d'une rampe de frequence pour mesurer la duree de propagation de signaux h.f. WO1996014590A1 (fr)

Applications Claiming Priority (4)

Application Number Priority Date Filing Date Title
CH3341/94-0 1994-11-08
CH334194 1994-11-08
CH368194 1994-12-05
CH3681/94-1 1994-12-05

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WO1996014590A1 true WO1996014590A1 (fr) 1996-05-17

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EP0863409A1 (fr) * 1997-03-04 1998-09-09 Thomson-Csf Procédé et dispositif de détection radar à modulation de fréquence à onde continue présentant une levée d'ambiguité entre la distance et la vitesse

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US4725841A (en) * 1983-06-30 1988-02-16 X-Cyte, Inc. System for interrogating a passive transponder carrying phase-encoded information
GB2160726A (en) * 1984-06-24 1985-12-24 Hewlett Packard Co Frequency synthesizer
US5374903A (en) * 1988-04-22 1994-12-20 Hughes Aircraft Company Generation of wideband linear frequency modulation signals
DE4244608A1 (de) * 1992-12-31 1994-07-07 Volkswagen Ag Computerisiertes Radarverfahren zur Messung von Abständen und Relativgeschwindigkeiten zwischen einem Fahrzeug und vor ihm befindlichen Hindernissen
US5345470A (en) * 1993-03-31 1994-09-06 Alexander Richard O Methods of minimizing the interference between many multiple FMCW radars

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Cited By (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
EP0863409A1 (fr) * 1997-03-04 1998-09-09 Thomson-Csf Procédé et dispositif de détection radar à modulation de fréquence à onde continue présentant une levée d'ambiguité entre la distance et la vitesse
FR2760536A1 (fr) * 1997-03-04 1998-09-11 Thomson Csf Procede et dispositif de detection radar a modulation de frequence a onde continue presentant une levee d'ambiguite entre la distance et la vitesse
US5963163A (en) * 1997-03-04 1999-10-05 Thomson-Csf Method and device for frequency-modulated continuous-wave radar detection with removal of ambiguity between distance and speed

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