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WO1993011582A1 - Antenne microruban compacte a bande large - Google Patents

Antenne microruban compacte a bande large Download PDF

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Publication number
WO1993011582A1
WO1993011582A1 PCT/US1992/009439 US9209439W WO9311582A1 WO 1993011582 A1 WO1993011582 A1 WO 1993011582A1 US 9209439 W US9209439 W US 9209439W WO 9311582 A1 WO9311582 A1 WO 9311582A1
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WO
WIPO (PCT)
Prior art keywords
antenna
spiral
radiation
microstrip
substrate
Prior art date
Application number
PCT/US1992/009439
Other languages
English (en)
Inventor
Johnson J. H. Wang
Victor K. Tripp
Original Assignee
Georgia Tech Research Corporation
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Georgia Tech Research Corporation filed Critical Georgia Tech Research Corporation
Priority to EP92925063A priority Critical patent/EP0614578A4/fr
Priority to JP5510113A priority patent/JPH07501432A/ja
Publication of WO1993011582A1 publication Critical patent/WO1993011582A1/fr

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Classifications

    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q1/00—Details of, or arrangements associated with, antennas
    • H01Q1/36—Structural form of radiating elements, e.g. cone, spiral, umbrella; Particular materials used therewith
    • H01Q1/38—Structural form of radiating elements, e.g. cone, spiral, umbrella; Particular materials used therewith formed by a conductive layer on an insulating support
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q21/00—Antenna arrays or systems
    • H01Q21/06—Arrays of individually energised antenna units similarly polarised and spaced apart
    • H01Q21/20—Arrays of individually energised antenna units similarly polarised and spaced apart the units being spaced along or adjacent to a curvilinear path
    • H01Q21/205—Arrays of individually energised antenna units similarly polarised and spaced apart the units being spaced along or adjacent to a curvilinear path providing an omnidirectional coverage
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q25/00—Antennas or antenna systems providing at least two radiating patterns
    • H01Q25/04—Multimode antennas
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q9/00—Electrically-short antennas having dimensions not more than twice the operating wavelength and consisting of conductive active radiating elements
    • H01Q9/04—Resonant antennas
    • H01Q9/16—Resonant antennas with feed intermediate between the extremities of the antenna, e.g. centre-fed dipole
    • H01Q9/26—Resonant antennas with feed intermediate between the extremities of the antenna, e.g. centre-fed dipole with folded element or elements, the folded parts being spaced apart a small fraction of operating wavelength
    • H01Q9/27—Spiral antennas

Definitions

  • the present invention relates generally to antennas, and more particularly relates to microstrip antennas.
  • FI antenna frequency-independent antenna
  • Such frequency-independent antennas typically have a radiating or driven element with spiral, or log-periodic structure that enables the frequency-independent antenna to transmit and receive signals over a wide band of frequencies, typically on the order of a 9:1 ratio or more (a bandwidth of 900%).
  • a lossy cylindrical cavity is positioned to one side of the antenna element so that when transmitting, energy effectively is radiated outwardly from the antenna only from one side of the antenna element (the energy radiating from the other side of the antenna element being dissipated in the cavity).
  • the antenna be mounted substantially flush with its exterior surface, in this case the skin of the aircraft.
  • This undesirably requires that the cavity portion of the frequency-independent antenna be mounted within the structure of the aircraft, necessitating that a substantial hole be formed therein to accommodate the cylindrical cavity, which typically is at least two inches deep and several inches in diameter.
  • the use of a lossy cavity to dissipate radiation causes about half of the radiated power to be lost, requiring a greater power input to effect a given level of power radiated outwardly from the frequency-independent antenna.
  • microstrip patch antenna In recent years the so-called "microstrip patch antenna" has been developed. See for example, U.S. Patent Reissue No. 29,911 of Munson (a reissue of U.S. Patent No. 3,921,177) and U.S. Patent Reissue No. 29,296 of Krutsinger, et al. (a reissue of U.S. Patent No. 3,810,183).
  • a typical microstrip patch antenna a thin metal patch, usually of circular or rectangular shape, is placed adjacent to a ground plane and is spaced a small distance therefrom by a dielectric spacer.
  • Microstrip patch antennas have generally suffered from having a narrow useful bandwidth, typically less than 10%.
  • 07/695,686 recites a multi-octave spiral-mode microstrip antenna which overcomes many of the prior art limitations.
  • This spiral-mode antenna approaches the bandwidth of frequency-independent antennas and is nearly flushly mounted above a ground plane.
  • multi-mode operation of a spiral-mode microstrip antenna requires the spiral to be of circumference at least m ⁇ , where m is the highest desired mode and ⁇ is the wavelength.
  • the spiral diameter can become undesirably large, especially at lower frequencies.
  • Microstrip patch array antennas have also been known in the art. See, for example, Munson, R.E., Conformal Microstrip Antennas and Microstrip Phased Arrays, IEEE Transactions on Antennas and Propogation, p. 74 (Jan. 1974).
  • the Munson article discusses an array of rectangular elements.
  • known microstrip arrays, including the Munson design generally are electrically large (i.e., the antenna is relatively large in comparison with the wavelength of the operating frequency), having individual elements of approximately one-half wavelength in diameter and spaced from one another a distance slightly greater than their diameters.
  • U.S. Patent No. 4,766,444 of Conroy et al relates to a conformal "cavity-less" antenna having an array of single-arm spiral elements driven in unison and which are aligned linearly along an outwardly-curved surface.
  • a lossy hex-cell structure spaces the spiral elements away from the ground plane and takes the place of the typical cavity.
  • the resulting antenna is disclosed as being suited for use as an interferometer and tends to suffer from having a narrow useful bandwidth. Again this is an electrically large array.
  • the present invention comprises a compact broadband microstrip antenna.
  • the invention comprises a microstrip structure for mounting to one side of a ground plane or other surface, the antenna comprising a closed (typically circular) array of antenna elements, each element positioned to one side of a substrate for spacing the elements a selected distance from the ground plane, the substrate having a low dielectric constant.
  • the elements are adapted to be electrically driven out of phase from one another to excite spiral modes.
  • the closed array comprises a circular arrangement of four or more elements, each element being made from a thin metal foil.
  • the substrate has a dielectric constant of between 1 and 4.5.
  • the thickness of the substrate is carefully selected to get near maximum gain at a particular wavelength, with the substrate having a thickness typically in the range of 0.1 to 0.30 inches for microwave frequencies of 2 to 18 GHz. The substrate thickness for other frequencies is determined by the frequency scaling method.
  • a loading material can be positioned adjacent the antenna elements.
  • an antenna which can be mounted externally to a structure and which can be conformed to the surface thereof. Also, the antenna exhibits a fairly broad bandwidth, typically on the order of 300%.
  • This design is based on the discovery by the applicants that the ground plane of a microstrip antenna is compatible with the spiral modes of the antenna.
  • the invention comprises a microstrip antenna for mounting to one side of a ground plane or other surface, the antenna comprising one or more antenna elements positioned to one side of a magnetic substrate for spacing the antenna elements a selected distance from the ground plane.
  • the magnetic substrate is chosen to have a relative permittivity which is roughly equal to its relative permeability. This allows the antenna to generate multiple spiral modes effectively, without the ill-effects of having a substrate with a high dielectric constant.
  • the present invention comprises a microstrip antenna for mounting to one side of a ground plane and includes one or more antenna elements positioned to one side of a substrate.
  • the present invention comprises a multioctave spiral-mode microstrip antenna system for mounting to one side of, or including, a ground surface.
  • the antenna system includes an antenna having a spiral-mode antenna element and a substrate positioned to one side of the anenna element for spacing the antenna element a selected distance from the ground surface, the selected distance being between 1/60 and 1/2 wavelengths throughout the multioctave operating frequency range.
  • the substrate has a relative dielectric constant of between 1.0 and 2.0, preferably as close to 1.0 as possible.
  • the antenna system also includes a feed network with a near-perfect impedance matching with the spiral mode antenna element to excite the desired spiral modes (with the effect of the ground plane incorporated in the impedance matching).
  • Fig. 1 is a plan view of a microstrip antenna in a preferred form of the invention.
  • Fig. 2A is a schematic, partially sectional side view of the antenna of Fig. 1.
  • Fig. 2B is a schematic, partially sectional side view of a portion of the antenna of Fig. 2A.
  • Fig. 3 is a schematic view of a feed for driving the antenna of Fig 1.
  • Figs. 4A and 4B are plan views of modified forms of the antenna of Fig. 1, depicting sinuous antenna elements.
  • Figs. 5A and 5B are plan views of modified forms of the antenna of Fig. 1, depicting log-periodic tooth antenna elements.
  • Fig. 6 is a plan view of a modified form of the antenna of Fig. 1, depicting a rectangular spiral antenna element.
  • Figs. 7 and 8 are plan views of modified forms of the antenna on Fig. 1, depicting Archimedean and equiangular spiral antenna elements, respectively.
  • Figs. 9A and 9B and 10A and 10B are schematic illustrations of mathematical models used to analyze the theoretical basis of the antenna of Fig. 1.
  • Figs. 11A and 11B are graphs of experimental laboratory results of the disruptive effect of the dielectric substrate (when the dielectric constant is great) on the radiation pattern of an antenna as shown in Fig. 1.
  • Fig. 12 is a graph of laboratory results comparing antennas according to the present invention with a prior cavity-loaded spiral antenna.
  • Fig. 13 is a graph of laboratory results for the antenna of Fig. 1 showing the effect of positioning the antenna element on antenna gain at various spacings from the ground plane for three different operating frequencies.
  • Fig. 15 is a schematic plan view of an antenna according to another preferred form and having closed array elements.
  • Fig. 16 is a side sectional view of the antenna of Fig. 15.
  • Fig. 18 is a schematic plan view of an alternative embodiment in which concentric circular arrays of elements are arranged.
  • Fig. 19 is a schematic illustration of a tunable multiple-resonance-frequency microstrip antenna switched by PIN diodes.
  • Fig. 20 is a schematic illustration showing that a substrate material used in a spiral-mode microstrip antenna with equal relative permittivity and permeability.
  • Figs. 21A and 21B show mode-2 antennas with a non-constant spacing above the ground plane.
  • Figs. 1, 2A and 2B show a multi-octave microstrip antenna 20, according to a preferred form of the invention and shown mounted to one side of a ground plane GP.
  • the antenna 20 includes an antenna element 21 comprising a very thin metal foil 21a, preferably copper foil, and a thin dielectric backing 21b.
  • the antenna element foil 21a shown in Figs. 1, 2A and 2B has a spiral shape or pattern including, first and second spiral arms 22 and 23. Spiral arms 22 and 23 originate at terminals 26 and 27 roughly at the center of antenna element 21.
  • the spiral arms 22 and 23 spiral outwardly from the terminals 26 and 27 about each other and terminate at tapered ends 28 and 29, thereby roughly defining a circle having a diameter D and a corresponding circumference of ⁇ D .
  • the antenna element foil 21a is formed from a thin metal foil or sheet of copper by any of well known means, such as by machining, stamping, chemical etching, etc.
  • Antenna element foil 21a has a thickness t of less than 10 mils or so, although other thicknesses obviously can be employed as long as it is thin in terms of the wavelength, say for example, 0.01 wavelength or less.
  • the antenna can be constructed to include its own ground plane, making the antenna suitable for mounting on non-conducting surfaces, e.g., on engineering plastics and composites.
  • the thin antenna element 21 is flexible enough to be mounted to generally nonplanar, contoured shapes of the ground plane, although in Figs. 2A and 2B the ground plane is represented as being truly planar.
  • the antenna element foil 21a is uniformly spaced a selected distance d (the standoff distance) from the ground plane GP by a dielectric spacer 32 positioned between the antenna element 21 and the ground plane GP.
  • the dielectric spacer 32 preferably has a low dielectric constant, in the range of 1 to 4.5, as will be discussed c more detail below.
  • the dielectric spacer 32 is generally in the form of a disk and is sized to be slightly smaller in diameter than the antenna element 21.
  • the thickness d of the dielectric spacer 32 typically is much greater than the thickness of the dielectric backing 21b of the antenna element 21.
  • the thickness d of spacer 32 typically is in the neighborhood of 0.25" for microwave frequencies.
  • the specific thickness chosen to provide a maximum gain for a given frequency should be no greater than one-half of the wavelength of the frequency in the medium of the dielectric spacer.
  • a loading 33 comprising a microwave absorbing material, such as carbon-impregnated foam, in the shape of a ring is positioned concentrically about dielectric spacer 32 and extends partially beneath antenna element 21.
  • a paint laden with carbon can be applied to the outer edge of the antenna element.
  • the antenna element can be provided with a peripheral shorting ring positioned adjacent and just outside the spiral arms 22 and 23 and the peripheral shorting ring (unshown) can be painted with the carbon-laden paint.
  • First and second coaxial cables 36 and 37 extend through an opening 38 in the ground plane GP for electrically coupling the antenna element 21 with a feed source, driver or detector.
  • the coax cables 36 and 37 include central shielded electric cables 42 and 43 which are respectively connected with the terminals 26 and 27.
  • the outer shieldine s of the coaxial cables 36 and 37 are electrically coupled to each other in the vicinity of the antenna element, as shown in Fig. 2B. As shown schematically in Fig. 3, this electrical coupling of the shielding of the coaxial cables can be accomplished by soldering a short electric cable 44 at its ends to each of the coaxial cables 36 and 37.
  • the coaxial cables 36 and 37 are connected to a conventional RF hybrid unit 46 which is in turn connected with a single coax cable input 47.
  • the function of the RF hybrid unit 46 is to take a signal carried on the input coax cable 47 and split it into two signals, with one of the signals being phase-shifted 180° relative to the other signal. The phase-shifted signals are then sent out through the coaxial cables 36 and 37 to the antenna element 21.
  • a balun may be used to split the input signal into first and second signals, with one of the signals being delayed relative to the other.
  • the dissipative loading 33 can be done away with by using a substrate with a very low relative dielectric constant, preferably close to unity (1.0), and by using a feed network with a near-perfect impedance match with the arms of the spiral to excite the desired spiral modes (with the effect of the ground plane incorporated in the impedance matching).
  • Fig. 4A shows an alternative embodiment of the antenna of Fig. 1, with the spiral arms 22 and 23 of Fig. 1 being replaced with sinuous arms 52 and 53. While a two-arm sinuous antenna element is shown in Fig. 4A, a four-arm sinuous antenna element can be provided if higher-order modes are desired, as shown in Fig. 4B.
  • Fig. 5A shows a modified form of the antenna element 21 in which the spiral arms 22 and 23 are replaced with log-periodic toothed arms 56 and 57.
  • the toothed antenna element illustratively shown in Fig. 5A includes toothed arms which have linear segments which are perpendicular to each other, i.e., the "teeth" of each arm are generally rectangular.
  • the teeth can be smoothly contoured to eliminate the sharp corners at each tooth.
  • the teeth can be curved as shown in Fig. 5B.
  • Fig. 6 shows another modified form of the antenna element of Fig. 1 in which the spiral arms 22 and 23 are replaced with rectangular spiral arms 58 and 59.
  • Each of the spiral arms is in the form of a spiraling square, as compared with the rounded spiral of the antenna element of Fig. 1.
  • Figs. 7 and 8 show that the spiral pattern of Fig. 1 can be provided as an "Archimedean spiral” as shown in Fig. 7 or as an "equiangular spiral” as shown in Fig. 8. 2. Theoretical Basis of the Mounting Arrangement
  • the basic planar spiral antenna which consists of a planar sheet of an infinitely large spiral structure, radiates on both sides of the spiral in a symmetric manner.
  • Figs. 9A and 9B depict an infinite, planar spiral backed by a ground plane.
  • the spiral mode fields in Region l can be decomposed into TE and TM fields in terms of vector potentials F l and A l as follows:
  • Equations (22) have six parameters in the five equations. Let, say A 1 , be given, then we can solve for all the other five parameters.
  • the spiral radiation modes can be supported by the structure of an infinite planar spiral backed by a ground plane as shown in Figure 1. This finding is the design basis of the multi-octave spiral-mode microstrip antennas disclosed herein.
  • the spiral is truncated.
  • the residual current on the spiral beyond the mode-1 active region therefore, faces a discontinuity where the energy is diffracted and reflected.
  • the diffracted and reflected power due to the truncation of the spiral, as well as possible mode impurity at the feed point, is believed to degrade the radiation pattern. Indeed, this is consistent with what we have observed.
  • Region 2 is an infinite dielectric medium with ⁇ 2 and ⁇ 0 .
  • the substrate thickness d is reduced to 1/32 inch, the effect of the dielectric becomes larger, especially at lower frequencies.
  • VSWR voltage standing-wave ratio
  • the spiral microstrip antenna is to be mounted on a curved surface.
  • a 3-inch diameter spiral microstrip antenna on a half-cylinder shell with a radius of 6 inches and a length of 14 inches.
  • the truncated spiral was placed 0.3-inch above and conformal to the surface of the cylinder with a styrofoam spacer.
  • a 0.5 inch-wide ring of microwave absorbing material was placed at the end of the truncated spiral, with half of the absorbing material lying inside the spiral region and half outside it.
  • the ring of absorbing material was 0.3-inch thick, thus filling the gap between the spiral antenna element and the cylinder surface.
  • the VSWR measurement of the spiral microstrip antenna conformally mounted on the half-cylinder shell was below 1.5 between 3.6 GHz and 12.0 GHz, and was below 2.0 between 2.8 GHz and 16.5 GHz. Thus, a 330% bandwidth was achieved for VSWR of 1.5 or lower, and a 590% bandwidth for VSWR of 2.0 or lower was reached.
  • the measured radiation patterns over ⁇ on the y-z principal plane with ⁇ -90° yielded good rotating-linear patterns obtained over a wide frequency bandwidth of 2-10 GHz.
  • the spiral-mode microstrip antenna can be conformally mounted on a curved surface with little degradation in performance for the range of radius of curvature studied here.
  • One technique for removing the residual power is to place a ring of absorbing material at the truncated edge of the spiral outside the radiation zone. This scheme allows the absorption of the residual power which would radiate in "negative" modes, which cause deterioration of the radiation patterns, especially their axial ratio. This scheme is shown in Figs. 1 and 2A by the provision of the loading ring 33.
  • Performance tests were conducted for a configuration similar to that shown in Figure 1, except that the spiral was Archimedean as shown in Fig. 7, with a separation between the arms of about 1.9 lines per inch.
  • the experimental results demonstrate that for a spacing d (standoff distance) of 0.145 inch, the impedance band is very broad -- more than 20:1 for a VSWR below 2:1.
  • the band ends depend on the inner and outer terminating radii of the spiral.
  • the feed was a broadband balun made from a 0.141 inch semi-rigid coaxial cable, which made a feed radius of 0.042 inch. It was necessary to create a narrow aperture in the ground plane in order to clear the balun.
  • the cavity's radius was 0.20 inch, and its depth 2 inches. This aperture also affects the high frequency performance.
  • each spiral (the Archimedean and the equiangular) was 3.0 inches, with foam absorbing material (loading) extending from 1.25 to 1.75 inches from center. If this terminating absorber is effective enough, the antenna match can be extended far below the frequencies at which the spiral radiates significantly. More importantly, at the operating frequencies, the termination eliminates currents that would be reflected from the outer edge of the spiral and disrupt the desired pattern and polarization. These reflected waves are sometimes called "negative modes" because they are mainly polarized in the opposite sense to the desired mode. Thus, their primary effect is to increase the axial ratio of the patterns.
  • the Archimedean and equiangular antennas operate well from 2 to 14 GHz, a 7:1 band. It is expected that the detailed engineering required to produce a commercial antenna would yield excellent performance over most of this range.
  • the gain is higher than that of a 2.5" commercial lossy-cavity spiral antenna up through 12 GHz, as shown in Figure 12. (We believe that the dip at 4 GHz is an anomaly.)
  • the increased gain of antennas of the present invention over a lossy-cavity spiral antenna is in part attributable to the relative lack of loss of radiated power from the underside of the spiral mode antenna elements.
  • the spiral mode antenna element radiates to both sides, with radiation from the underside passing through the dielectric backing and the dielectric substrate relatively undiminished. This radiation is reflected by the ground plane (sometimes more than once) and augments the radiation emanating from the upper side.
  • Fig. 12 also shows gain curves for a ground plane spacing of 0.3 inch.
  • the Archimedean version of this design demonstrates a gain improvement over the nominal loaded-cavity level of 4 .5 dBi (with matched polarization) over a 5:1 band.
  • the gain of the 0.145 inch spaced antenna is lower because the substrate was a somewhat lossy cardboard material rather than a light foam used for the 0.3 inch example.
  • FIG. 13 shows gain plotted at several frequencies as a function of spacing for a "substrate" of air. At low frequencies, the spiral arms act more like transmission lines than radiators as they are moved closer to the ground plane. They carry much of their energy into the absorber ring, and the gain decreases.
  • edge loading most notably foam absorbing material and magnetic RAM (radar absorbing materials) materials.
  • foam absorbing material we compared log-spirals terminated with a simple circular truncation (open circuit) and terminated with a thin circular shorting ring. There was no discernable difference in performance.
  • the magnetic RAM absorber was tried on open-circuit Archimedean and log-spirals with spacings of 0.09 and 0.3 inches. The results show that the magnetic RAM is not nearly so well-behaved as the foam. In addition to the gain loss caused by the VSWR spikes, the patterns showed a generally poor axial ratio, indicating that the magnetic RAM did not absorb as well as the foam.
  • the loading materials were always shaped into a one-half-inch wide annulus, half within and half outside the spiral edge. The thickness was trimmed to fit between the spiral and the ground plane, and in the very close configurations it was mounted on top of the spiral.
  • This disclosure presents an analysis, supported by experiments, of a multi-octave, frequency-independent or spiral-mode microstrip antenna according to the present invention. It shows that the spiral-mode structure is compatible with a ground plane backing, and thus explains why and how the spiral-mode microstrip antenna works.
  • Spiral modes refers to eigenmodes of radiation patterns for structures such as spiral and sinuous antennas. Indeed, each of the spiral, sinuous, log-periodic tooth, and rectangular spiral antenna elements disclosed herein as examples of the present invention exhibit spiral modes.
  • the axis perpendicular to the plane of the antenna points to zero degrees in the figure.
  • the "spiral mode" antenna elements disclosed herein as part of a microstrip antenna radiate in patterns roughly similar to, though not necessarily identical with, the patterns of Fig. 14.
  • Multioctave refers to a bandwidth of greater than 100%.
  • Frequency-independent refers to a geometry characterized by angles or a combination of angles and a logarithmically periodic dimension (excepting truncated portions), as described in R.H. Rumsey in Frequency Independent Antennas, supra.
  • the stand-off distance d should be between 0.015 and 0.30 of a wavelength of the waveform in the substrate (the dielectric spacer).
  • the relative dielectric constant of the substrate applicants have found that materials with ⁇ of between 1 and 4.37 work well, and that a range of 1.1 to 2.5 appears practical. A higher dielectric constant (5 to 20) leads to gradual narrowing of bandwidth and deterioration of performance which nevertheless may still be acceptable in many applications.
  • This and other design configurations, which operate satisfactorily for a specific frequency range, can be changed so that the antenna will work satisfactorily in another frequency range of operation. In such cases the dimensions and dielectric constant of the design are changed by the well known "frequency scaling" technique in antenna theory.
  • an antenna 60 is mounted above a ground plane GP and includes a somewhat stiff, comformable backing 61.
  • the backing 61 is a unitary structure, preferably made of printed circuit board material.
  • the backing 61 is spaced above the ground plane GP by a dielectric spacer 62 in accordance with the principles set forth in the above numbered sections 1-3.
  • a closed array or series of patch elements 63, 64, 65, 66, 67, 68, 69, and 70, is formed atop the upper surface of the backing 61 by conventional techniques, such as by photoetching.
  • the array is circular, although what is essential is that the array be "closed", i.e., is generally of the form of a loop. While eight elements are depicted in Fig. 15, a greater or lesser number of elements can be used. In Fig. 16, the vertical dimensions of the patch elements and of the backing are exaggerated somewhat to make these elements more visually discernible in the figure.
  • the patch elements 63-70 are connected to unshown electrical means for driving the individual elements, the driving means being adapted to drive the individual patch elements in a phased manner.
  • the electrical circuitry used to phase signals delivered to the individual patch elements is well-known.
  • the signal is split up into several signals and delayed or phase-shifted an appropriate amount, by a network of "hybrids” sometimes called a “processor”, before being delivered to the patch elements.
  • a network of “hybrids” sometimes called a “processor”
  • the individual patch elements 63-70 are electrically coupled with the driving means in a manner similar to that shown in Fig. 2B, i.e., through the use of cabling or in another suitable manner.
  • the structure just described is extremely compact and is well-suited for being used on the surface of an object, for example, on the surface of an airplane.
  • the antenna 60 with the array of individual antenna elements 63-70 has a small overall dimension for a bandwidth of 30 to 300%, depending on the diameter of the array.
  • This arrangement allows the antenna to be made substantially smaller than prior antennas at a sacrifice of some bandwidth and some gain, and that the smaller the diameter of the circular array, the smaller the bandwidth.
  • the present invention allows the diameter of the antenna to be reduced by up to 2/3 or so.
  • the spiral-mode circular array fills the need for a conformable, low-profile, antenna with a moderately wide bandwidth in the 30% to 300% range while the array diameter can be only 1/2 to 1/3 the spiral diameter.
  • the basic concept of a spiral-mode circular phased array is shown in Figure 15.
  • the circular array is on a x-y plane which is treated as a horizontal plane parallel to the earth.
  • the array elements are on a circle of radius a, and can be represented as either magnetic or electric current elements, denoted by J m n for the nth element of mode m.
  • the current J m n must have a polarization, amplitude, and phase as follows:
  • this circular array can provide the spatial coverage shown in Figures 17A and 17B. Now if two or more of these modes are combined, the resultant pattern has a narrower steerable beam, as well as one or more steerable nulls for noise or interference reduction.
  • This multi-mone circular array alternatively can be realized, as in the co-pending patent application, by a multimode planar spiral, for which the radiation current band theory is well known.
  • the planar spiral requires a much larger aperture, because its radiation occurs on a circle whose circumference is m ⁇ in length.
  • the planar spiral can be unattractively large.
  • the array radius a can be arbitrarily small.
  • the tolerance of the array becomes increasingly stringent as the array diameter is reduced to below about 0.3 ⁇ for mode 1 and 0.6 ⁇ for mode 2.
  • a major advantage of this spiral-mode circular array is its ability to radiate, especially for higher-order modes (m > 2), on a smaller aperture.
  • a planar spiral needs to have a circumference of more than 3 ⁇ (a diameter of 0.955 ⁇ ).
  • a ⁇ circumference (0.318 ⁇ in diameter) is acceptable.
  • the tolerance requirements on the feed network becomes more and more stringent for smaller apertures.
  • the individual microstrip patch antenna is known for its narrow bandwidth, typically 10% and often 3 to 6%.
  • the bandwidth of a microstrip antenna can be increased.
  • the bandwidth at 10 GHZ is about 20%.
  • the impedance bandwidth of the array can be made much larger than that of the individual array elements.
  • the signal-to-noise ratio of the antenna disclosed herein should be equivalent to the single-element low-gain antennas with broad apple or doughnut beams.
  • PIN diodes As shown in Fig. 19. This technique of switching the effective Iength of a microstrip structure has been experimentally investigated and analyzed in some instances. The high temperature limits for this diode-switching device are yet to be determined.
  • the substrate between the antenna element and the ground plane is a magnetic material with equal relative permittivity and permeability
  • the spiral modes can radiate effectively.
  • a substrate having high relative permittivity say, greater than 5
  • the antenna pattern begins to deteriorate.
  • the substrate is compatible with the spiral modes and therefore good radiation patterns for each mode can be generated without other unwanted modes that can disrupt the pattern.
  • antenna element (s) 72 is positioned atop a magnetic substrate 73 having substantially equal relative permittivity and permeability.
  • a loading material 74 is placed about the periphery.
  • the relative permittivity and permeability of the magnetic substrate are chosen to be a higher number, say, 10, then the wavelength in the substrate will be only 1/10 (10%) of that in free space. This allows the antenna size to be reduced to 1/10 (one-tenth) of its size when using a honey-comb substrate (relative permittivity and permeability being close to unity).
  • Fig. 20 shows that a magnetic material is used as the substrate 73 for the spiral-mode microstrip antenna.
  • ⁇ r since ⁇ r ⁇ r .
  • custom materials can be constructed by mixing grains of two materials to achieve equal, or nearly equal, relative permittivity and permeability.
  • the size of the grains must be small in comparison with wavelength (in the material), and must be uniformly distributed to achieve homogeneity on a macroscopic scale.
  • two different types of cubes, one more dielectric and the other more magnetic, and with their linear dimensions being identically equal to 0.1 wavelength (in the material) can be alternately spaced to approximate a homogeneous material of equal relative permittivity and permeability.
  • Another method of making custom magnetic material for substrate of equal ⁇ r and ⁇ r is to place electrically thin dielectric and magnetic sheets parallel to the ground plane alternately in a stack. (Sheets placed perpendicular to the ground place should have similar effects.) The stack then appears macroscopically to be homogeneous with equal ⁇ r and ⁇ r .
  • the physical size of a rnode-2 antenna which generally has a larger and more complex feed network, can be reduced by varying the effective thickness of the substrate.
  • a simple coax feed at the center excites a transmission-line wave propagating away from the center along the spiral structure, thereby forming spiral modes.
  • this mode-2 antenna is not only a reduction in physical size, including that of its feed, but also a reduction in cost, improvement in reliability and greater structural simplicity.
  • the gain of the spiral-mode microstrip antenna drops sharply when the spacing between the antenna element and the ground plane is decreased to below, say, 0.02 wavelength. This phenomenon is taken advantage of in the following mode-2 antenna.
  • Figs. 21A and 21B show two versions of a simple illustrative design in which the center conductor of a coaxial line 76 is fed through a ground plane GP to the center of a spiral structure 77.
  • the two spiral arms within the mode-1 radiation region join at the center with the center conductor of the coaxial line.
  • the fine Archimedean spiral arms as shown in the mode-2 region are broadened in the mode-1 region.
  • the specific pattern of the broadening of the arms is not critical as long as it transforms the impedance (usually 50 ohms of the coax cable at the center into the impedance of the spiral microstrip structure.
  • Radiation in the mode-1 region is minimized by choosing d 1 , the spacing between the spiral element 77 and the ground plane, to be electrically small (less than say, 0.02 wavelength).
  • the wave moves outwardly from the center of the spiral structure and enters the mode-2 region (where the circumference is greater than about 1.1 wavelength)
  • effective radiation takes place because the spacing d 2 between the spiral element 77 and the ground plane GP is now greater than about 0.05 wavelength.
  • the fact that the radiation occurs in the mode-2 region means that the radiation pattern should be that of mode-2.
  • the spacing between the spiral element 77 and the ground plane abruptly changes from d 1 in the mode-1 region to d 2 in the mode-2 region.
  • radiation in mode-2 is effective.
  • the abrupt increase in spacing for substrate thickness from d 1 to d 2 causes undesired reflections.
  • the reflection between mode-1 and mode-2 regions is reduced by employing a tapered section to effect a gradual increase in substrate thickness from d 1 to d 2 .
  • the mode-2 radiation is not as effective at frequencies at which mode-2 regions begins in the tapered transition region, since the smaller substrate thickness in the transition region suppresses radiation.
  • the taper between d 1 and d 2 shown in Fig. 21B can be linear or of some other smooth curve, the selection of which is a tradeoff among several considerations, including technical performance as well as production cost and ruggedness.
  • ground plane is a large conducting sphere and the spiral is positioned outside it.
  • the patch elements can comprise lossy components for impedance matching.

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  • Waveguide Aerials (AREA)
  • Variable-Direction Aerials And Aerial Arrays (AREA)

Abstract

L'invention décrit des dispositifs d'antennes microruban compactes à bande large. Chaque dispositif comprend un matériau diélectrique (32, 62) sur un des côtés duquel sont montées les antennes microruban et comportant un plan de masse (GP) sur le côté opposé. Un premier dispositif comprend un groupement fermé d'éléments d'antenne (63-70) déphasés électriquement les uns des autres, de façon à exciter un ou plusieurs modes spiraux. Un deuxième dispositif comprend un ou plusieurs éléments d'antenne disposés sur un substrat magnétique (73) possédant une permittivité relative approximativement égale à sa perméabilité relative. Dans un troisième dispositif, l'antenne microruban fonctionne en monomode et le rayonnement provenant d'autres modes est supprimé au moyen de la modification de l'espacement situé au-dessus du plan de masse (GP) dans les zones de rayonnement, de façon à ne favoriser le rayonnement que dans les zones souhaitées. Dans un quatrième dispositif, un élément d'antenne en mode spiral (21) est éloigné du plan de masse (GP) en fonction d'une valeur située entre 1/60 et 1/2 de longueurs d'onde sur une plage de fréquence fonctionnant en multioctave. Le substrat (32) possède une constante diélectrique relative située à proximité de 1,0 et l'antenne est excitée par un réseau d'alimentation à adaptation d'impédance presque parfaite.
PCT/US1992/009439 1991-11-26 1992-11-04 Antenne microruban compacte a bande large WO1993011582A1 (fr)

Priority Applications (2)

Application Number Priority Date Filing Date Title
EP92925063A EP0614578A4 (fr) 1991-11-26 1992-11-04 Antenne microruban compacte a bande large.
JP5510113A JPH07501432A (ja) 1991-11-26 1992-11-04 小型広帯域マイクロストリップアンテナ

Applications Claiming Priority (2)

Application Number Priority Date Filing Date Title
US79870091A 1991-11-26 1991-11-26
US07/798,700 1991-11-26

Publications (1)

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WO1993011582A1 true WO1993011582A1 (fr) 1993-06-10

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EP (1) EP0614578A4 (fr)
JP (1) JPH07501432A (fr)
CA (1) CA2124459C (fr)
MX (1) MX9206637A (fr)
TW (1) TW222353B (fr)
WO (1) WO1993011582A1 (fr)

Cited By (6)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US5589842A (en) * 1991-05-03 1996-12-31 Georgia Tech Research Corporation Compact microstrip antenna with magnetic substrate
US5691734A (en) * 1994-06-01 1997-11-25 Alan Dick & Company Limited Dual polarizating antennae
US6400322B2 (en) 2000-04-07 2002-06-04 Industrial Technology Research Institute Microstrip antenna
GB2431049A (en) * 2005-10-05 2007-04-11 Motorola Inc Antenna arrangement
SE2030349A1 (en) * 2020-11-30 2021-12-21 Gapwaves Ab Improved ultra-wideband circular-polarized radiation element with integrated feeding
US20230074075A1 (en) * 2021-09-08 2023-03-09 MeshPlusPlus, Inc. Phased-array antenna with precise electrical steering for mesh network applications

Families Citing this family (3)

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Publication number Priority date Publication date Assignee Title
US7545338B2 (en) * 2006-11-16 2009-06-09 Tdk Corporation Log-periodic dipole array (LPDA) antenna and method of making
US8106849B2 (en) * 2009-08-28 2012-01-31 SVR Inventions, Inc. Planar antenna array and article of manufacture using same
CN107091847B (zh) * 2017-06-01 2023-11-07 厦门大学 一种介质材料电磁参数测量装置及测量方法

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US4651159A (en) * 1984-02-13 1987-03-17 University Of Queensland Microstrip antenna
JPH02246504A (ja) * 1989-03-20 1990-10-02 Fujitsu Ltd 移動体衛星通信システム用移動局アンテナ装置
US4962383A (en) * 1984-11-08 1990-10-09 Allied-Signal Inc. Low profile array antenna system with independent multibeam control
EP0394960A1 (fr) * 1989-04-26 1990-10-31 Kokusai Denshin Denwa Co., Ltd Antenne à microruban

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US4651159A (en) * 1984-02-13 1987-03-17 University Of Queensland Microstrip antenna
US4962383A (en) * 1984-11-08 1990-10-09 Allied-Signal Inc. Low profile array antenna system with independent multibeam control
JPH02246504A (ja) * 1989-03-20 1990-10-02 Fujitsu Ltd 移動体衛星通信システム用移動局アンテナ装置
EP0394960A1 (fr) * 1989-04-26 1990-10-31 Kokusai Denshin Denwa Co., Ltd Antenne à microruban

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IEEE Transaction on Antennas & Propagation, Volume 39, No. 3, March 1991, J.J.H. WANG et al., "Design of Multioctave Spiral-Mode Microstrip Antennas", pp. 332-335 *
IEEE Transactions on Antennas & Propagation, Volume 29, No. 1, January 1981, K.R. CARVER et al., "Microstrip Antenna Technology", pp. 2-23 *
See also references of EP0614578A4 *

Cited By (9)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US5589842A (en) * 1991-05-03 1996-12-31 Georgia Tech Research Corporation Compact microstrip antenna with magnetic substrate
US5691734A (en) * 1994-06-01 1997-11-25 Alan Dick & Company Limited Dual polarizating antennae
US6400322B2 (en) 2000-04-07 2002-06-04 Industrial Technology Research Institute Microstrip antenna
GB2431049A (en) * 2005-10-05 2007-04-11 Motorola Inc Antenna arrangement
GB2431049B (en) * 2005-10-05 2008-02-27 Motorola Inc An antenna arrangement
SE2030349A1 (en) * 2020-11-30 2021-12-21 Gapwaves Ab Improved ultra-wideband circular-polarized radiation element with integrated feeding
SE544087C2 (en) * 2020-11-30 2021-12-21 Gapwaves Ab Improved ultra-wideband circular-polarized radiation element with integrated feeding
US20230074075A1 (en) * 2021-09-08 2023-03-09 MeshPlusPlus, Inc. Phased-array antenna with precise electrical steering for mesh network applications
US12149005B2 (en) * 2021-09-08 2024-11-19 MeshPlusPlus, Inc. Phased-array antenna with precise electrical steering for mesh network applications

Also Published As

Publication number Publication date
JPH07501432A (ja) 1995-02-09
EP0614578A4 (fr) 1995-05-10
CA2124459A1 (fr) 1993-06-10
TW222353B (fr) 1994-04-11
MX9206637A (es) 1993-11-01
EP0614578A1 (fr) 1994-09-14
CA2124459C (fr) 1998-09-22

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