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WO1991002421A1 - Procede et dispositif pour la conversion de signaux de reception a modulation numerique haute frequence - Google Patents

Procede et dispositif pour la conversion de signaux de reception a modulation numerique haute frequence Download PDF

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Publication number
WO1991002421A1
WO1991002421A1 PCT/EP1990/001305 EP9001305W WO9102421A1 WO 1991002421 A1 WO1991002421 A1 WO 1991002421A1 EP 9001305 W EP9001305 W EP 9001305W WO 9102421 A1 WO9102421 A1 WO 9102421A1
Authority
WO
WIPO (PCT)
Prior art keywords
received signal
phase
signal
data
frequency
Prior art date
Application number
PCT/EP1990/001305
Other languages
German (de)
English (en)
Inventor
Gerhard Schultes
Arpad Ludwig Scholtz
Ernst Bonek
Original Assignee
SIEMENS AKTIENGESELLSCHAFT öSTERREICH
Siemens Aktiengesellschaft
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by SIEMENS AKTIENGESELLSCHAFT öSTERREICH, Siemens Aktiengesellschaft filed Critical SIEMENS AKTIENGESELLSCHAFT öSTERREICH
Priority to EP90912008A priority Critical patent/EP0486554B1/fr
Priority to US07/834,551 priority patent/US5402449A/en
Priority to DE59007578T priority patent/DE59007578D1/de
Publication of WO1991002421A1 publication Critical patent/WO1991002421A1/fr

Links

Classifications

    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/18—Phase-modulated carrier systems, i.e. using phase-shift keying
    • H04L27/22—Demodulator circuits; Receiver circuits
    • H04L27/233—Demodulator circuits; Receiver circuits using non-coherent demodulation
    • H04L27/2332—Demodulator circuits; Receiver circuits using non-coherent demodulation using a non-coherent carrier
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/0014—Carrier regulation
    • H04L2027/0024—Carrier regulation at the receiver end
    • H04L2027/0026—Correction of carrier offset
    • H04L2027/0028—Correction of carrier offset at passband only
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/0014—Carrier regulation
    • H04L2027/0024—Carrier regulation at the receiver end
    • H04L2027/0026—Correction of carrier offset
    • H04L2027/003—Correction of carrier offset at baseband only
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/0014—Carrier regulation
    • H04L2027/0044—Control loops for carrier regulation
    • H04L2027/0053—Closed loops
    • H04L2027/0057—Closed loops quadrature phase

Definitions

  • a reception method and a receiver for cordless telephones are described in Telcom Report 10 (1987), No. 2, pages 130 to 137.
  • the received signal is mixed into the baseband via a high frequency stage and several intermediate frequency stages.
  • the de odulation is carried out with a PLL circuit.
  • Linear modulation methods are divided into two-stage methods with binary amplitude modulation (BAM) and binary phase shift keying (BPSK), four-stage methods with quadrature amplitude modulation (QAM) and quadrature phase shift keying (QPSK), as well as multi-stage modifications.
  • FSK or frequency modulation-like methods such as minimum shift keying (MSK) and Gaussian-filtered minimum shift keying (GMSK), are customary as nonlinear modulation methods.
  • MSK minimum shift keying
  • GMSK Gaussian-filtered minimum shift keying
  • Demodulation is carried out for nonlinear modulation methods using PLL, Costas loop or discriminators. Synchronous quadrature demodulators are used for linear modulation methods.
  • the invention has for its object to enable the processing of two- and multi-stage digitally modulated signals with a receiver.
  • Baseband is converted and broken down into a real and imaginary part that is continuous in terms of time and amplitude, the phase difference per bit duration being kept smaller than half the phase shift caused by the modulation per bit due to the incoherent conversion, the signal parts subsequently being amplified, filtered and are sampled, the sampling time being determined within a preceding synchronization time, then the sampling signals being digitized and the sampling values being converted as a binary pair of numbers for the magnitude and angle of the vector, the magnitude value being used for gain control and from the difference between two successive angle values Classification received data can be recovered.
  • Image frequency avoided.
  • the incoherent demodulation avoids complex carrier synchronization and permits DC voltage-free signal paths.
  • the phase difference is kept low by frequency correction or oscillators with exact frequency constancy.
  • almost all two-stage and multi-stage, linear and non-linear modulation methods can be used with this reception method.
  • Efficient decoding of angles beyond 360 ° is achieved in that an angular value is buffered over a bit period, subtracted from the following in two's complement and the received data is recovered from a classification of the result which is dependent on the modulation method used.
  • the adaptation to the individual modulation process is done via a programmable setting of the classification, which means that no circuit change is necessary.
  • the sampling time for the angular value of a linearly modulated received signal is determined within a synchronization time before the transmission of valid data by changing the course of the amount of the received signal modulated with a synchronization sequence and after conversion to the baseband the sign function is formed and this is digitally differentiated, a zero sequence being formed, the temporal position of the pulses relative to the system clock being measured and averaged over several bits.
  • a correction of the phase difference by incoherent conversion is derived from the rotation of the vector of the received signal.
  • the sampling time for the angular value of a nonlinear modulated received signal is determined within a synchronization time before the transmission of valid data by over-sampling the carrier modulated with the synchronization sequence after conversion to the baseband and obtaining a phase shift therefrom , a phase difference being formed from this by forming the difference over a bit duration, the phase sign function of which is compared in phase with the data clock and the phase shift is averaged over several bits.
  • a receiver for digitally modulated signals in mobile communication systems for carrying out the method according to the invention is characterized in that an antenna is connected to a mixer for the formation of the real and imaginary part of the received signal, that the mixer has two low-pass filters adjustable amplification are arranged on the output side, each of which is connected via an analog / digital converter to a corrector for the magnitude and angle of the vector, the converter for the recovery of data from the recorder being connected to a decoder which contains a classifier and an adjusting device - device " for determining the sampling time is connected to the corrector and the decoder ' .
  • the implementation in the baseband with a mixer saves high-frequency circuitry.
  • the circuit components can be fully integrated in Si bipolar or MOS technology. There are no high * frequency or intermediate frequency levels can be adjusted more.
  • the control behavior of the digital amplitude control can be programmed and changed without changing the circuit.
  • the low-pass filter removes unwanted signal components.
  • the mixer stage is designed as a quadrature demodulator, which consists of two mixers which are connected to a local oscillator for feeding with an oscillator signal shifted by 0 "and 90" and the local oscillator has a retuning input which is connected to an offset correction device.
  • the local oscillator is not phase-locked to the input signal, which saves circuitry for synchronization.
  • the carrier frequency offset from the phase shift of the incoherence of the frequency of the input signal and local oscillator is minimized via the retuning input. Its value results from the determination of the sampling time and no longer has to be calculated additionally.
  • the amplification factor of the low-pass filter can be adjusted digitally and the corrector for this regulation is fed back to the low-pass filter.
  • the low-pass filters contain DC voltage separating capacitors. They therefore act like bandpass filters with a width of several decades.
  • the corrector consists of two permanent memories, the data addresses of which are formed by the numerical values for the real and imaginary part of the samples and that the absolute value and the angular value are present at the data output of the respective permanent memory.
  • the mixing stage In order to be able to extend the demodulation of the received signal to extremely high frequencies, it is advantageous for the mixing stage to be preceded by at least one intermediate frequency converter. is.
  • the field of application of the receiver is expanded in that it can be adapted to different reception frequencies by means of different intermediate frequency converters.
  • FIG. 1 is a block diagram of the first embodiment
  • FIG. 2 shows the time structure of the received signal
  • FIG. 4 shows the signal curve for determining the sampling time in the case of linear modulation methods
  • FIG. 5 shows a block diagram of the second exemplary embodiment.
  • FIG. 1 The block diagram of a receiver for two- and four-stage digitally modulated signals for use in mobile communication systems is shown in FIG. 1.
  • the first exemplary embodiment is explained with reference to this figure, with an inserted explanation of FIGS. 2-4, and with tables. It is a homodyne, incoherent, digital vector receiver with a mixing stage, analogs signal paths are indicated as a simple digi ⁇ tale data paths as' double line. The control branches are dashed accordingly.
  • a preamplifier VV connected to an antenna A raises the input level with little noise and intermodulation and at the same time decouples antenna A from the inputs of the mixer.
  • This high-frequency pre-stage contains only a fixed, rough pre-selection of the input frequency range.
  • the pre-amplifier VV is equipped with a controllable amplification to enable partial input dynamic compensation.
  • the input signal is converted in one step from the reception frequency f.-, around 2 GHz, to a baseband.
  • the reception signal can be broken down into real and imaginary parts. This is done by means of a quadrature demodulator, which consists of a power divider LT and two identical mixers M, which are fed with a 0 * and a 90 "phase-shifted signal from a local oscillator VCO with an oscillator frequency f. Q.
  • the local oscillator VCO is not frequency and phase locked to the input carrier frequency.
  • a retuning input is provided for a rough frequency correction at the local oscillator VCO.
  • the retuning voltage is determined in an offset correction device OKE.
  • G should not be more than 5 ppm.
  • Two linear low-pass filters TP effect the main selection of the receiver. They contain DC voltage isolating capacitors to suppress DC voltage components of the useful signal and therefore act like broadband bandpass filters.
  • the gain of the individual filter stages can be adjusted digitally, so that dynamic compensation can also be carried out here. Behind the filter stages are the real and imaginary parts of the received signal in band-limited, amplitude and time continuous form.
  • the signals in the real and imaginary branches are periodically sampled by sample-and-hold elements SH with a system clock ST and digitized in two analog / digital converters AD. Due to the dynamic compensation at the preamplifier VV and at the low-pass filters TP, the dynamic fluctuations are limited to a few dB and a resolution of the analog / digital converter AD of 5-8 bits is sufficient.
  • the system clock ST is generated by a clock generator TG and has a frequency of 8.8 MHz. Thereafter, the real and imaginary parts of the input signal are available as binary numerical values at all times.
  • the data rate is 1.1 Mbit / sec with a data clock DT of 1.1 MHz, with which an 8-fold oversampling is achieved by the system clock ST.
  • the conversion of the real and imaginary part into the amount and angle of a complex vector is carried out using a conversion table in the permanent memory ROM.
  • the numerical values for the real and imaginary part form the data addresses of the permanent memory ROM, the absolute value B and the angular value W are obtained from the data output.
  • the amount B is used to set the amplification in the preamplifier VV and in the low-pass filters TP of the main selection.
  • the angle information is coded such that N bits with a value set of 2 correspond to the full angle of 360 ° (see Table 1). This coding allows binary angle addition and subtraction with the correct sign, with two's complement beyond 360 °.
  • a subtractor SB forms the difference between the current angle value W and the angle value W of the previous sampling time stored in a buffer ZS.
  • a classification of the amount depending on the selected modulation method The angular difference ⁇ ⁇ in decision areas E (Table 2) then delivers demodulated receive data D, whose sampling time is determined in an adjusting device JE.
  • a classifier K and a latch L are connected to the subtractor SB to form a decoder DC. The value is selected from the latch L.
  • the vector receiver can be automatically converted to different modulation methods by a post-microcomputer. This enables the construction of a multi-standard device.
  • Tab. 2 Decision area of the vector receiver for two- and four-stage modulation methods
  • phase shift resulting from the incoherence of the input signal and local oscillator VCO per bit must be smaller than half the angle of a decision area E. From this follows the maximum permissible frequency difference f n U j FifaX between the received signal and the local oscillator signal
  • ⁇ F means the angle of a decision area E and T ... the bit duration, which is the reciprocal of the data DT.
  • the carrier frequency offset Af is made from the difference in the phase shifts
  • a frequency correction signal is derived as the tuning voltage.
  • a synchronization sequence for determining the sampling time must be sent at the beginning of each reception sequence ES. This is initiated by briefly blanking out the transmission signal. This results in the time structure of the received signal with blanking time AZ, synchronization time SZ and bit transmission time BZ. The time at which the sampling time is then valid is given by the stability of the transmission and reception data clock, and the fineness of the time resolution " for the sampling within a bit. The number N of validly received bits results as
  • phase shift PHD increases continuously as a result of the carrier frequency offset.
  • a phase difference PD can be formed by 8-times oversampling and phase difference formation with the value 1 bit previous at each sampling time within a bit.
  • Their angle sign function WVF corresponds to the synchronization sequence SF shifted by T ... / 2.
  • modulation with the synchronization sequence SF can be used to generate a zero-crossing of the received signal at periodic intervals, as shown in FIG. 4 for linear amplitude modulation with suppressed carrier.
  • the course of the carrier frequency offset ⁇ ⁇ f and the course of the real and imaginary part RT, IT of the received signal are shown.
  • the amount of the vector is variable with linear modulation methods.
  • narrow pulses are generated by means of digital differentiation at the time of the zero crossing. They form a zero pulse sequence NF.
  • the automatic gain control is a controller R implemented in digital technology, shown in Fig. 1. It has the task keep the fluctuating signal level constant during operation with a medium modulation of the analog / digital converter AD to a few dB.
  • the absolute value B formed in the permanent memory ROM is read into the controller R as the actual level.
  • the controller R must work so slowly that it cannot follow the brief level drops before the synchronization times.
  • the non-band-limited signal power present after the mixers M is used as a parameter for distributing the amplification to the RF pre-stage and the main selection stages.
  • the parameter is determined by forming the amount of the reception vector in a normalization device NV.
  • FIG. 5 shows the schematic block diagram of a heterodyne vector receiver as a second exemplary embodiment. It works like the vector receiver of the first embodiment with incoherent demodulation.
  • the signal received via antenna A is reduced from 60 GHz to 2 GHz via an intermediate frequency converter ZFU.
  • a vector receiver which is constructed as in the first exemplary embodiment, can be used in module technology for the reception of satellite signals.
  • the processing of the received signal from the preamplifier VV remains the same.
  • the intermediate frequency converter ZFU contains an input amplifier EV, which is followed by a premixer VM.
  • the premixer VM is connected to an intermediate frequency oscillator ZFO.

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  • Engineering & Computer Science (AREA)
  • Computer Networks & Wireless Communication (AREA)
  • Signal Processing (AREA)
  • Digital Transmission Methods That Use Modulated Carrier Waves (AREA)
  • Radar Systems Or Details Thereof (AREA)
  • Superheterodyne Receivers (AREA)
  • Folding Of Thin Sheet-Like Materials, Special Discharging Devices, And Others (AREA)
  • Pile Receivers (AREA)

Abstract

Sur un récepteur de signaux à modulation numérique dans des systèmes de communication mobiles, le signal de réception peut être représenté sous la forme d'un vecteur complexe. Le signal fait l'objet d'une conversion incohérente homodyne ou hétérodyne dans la bande de base par l'intermédiaire d'un étage de mélange comportant des mélangeurs (M). On effectue une décomposition en une partie réelle et une partie imaginaire. Les parties de signal sont filtrées par des filtres passe-bas (TP) et numérisées au moyen de convertisseurs analogique-numérique (AD). Les valeurs lues sont transformées sous forme de valeur et d'angle de vecteur. La valeur (B) règle l'amplification du préamplificateur (VV) et des filtres passe-bas (TP) et les données de réception (D) sont récupérées à partir de la différence de deux valeurs d'angle (W) successives.
PCT/EP1990/001305 1989-08-11 1990-08-08 Procede et dispositif pour la conversion de signaux de reception a modulation numerique haute frequence WO1991002421A1 (fr)

Priority Applications (3)

Application Number Priority Date Filing Date Title
EP90912008A EP0486554B1 (fr) 1989-08-11 1990-08-08 Procede et dispositif pour la conversion de signaux de reception a modulation numerique haute frequence
US07/834,551 US5402449A (en) 1989-08-11 1990-08-08 Process and device for converting digitally modulate high-frequency reception signals
DE59007578T DE59007578D1 (de) 1989-08-11 1990-08-08 Verfahren und vorrichtung zum umsetzen digital modulierter empfangssignale aus dem hochfrequenzbereich.

Applications Claiming Priority (2)

Application Number Priority Date Filing Date Title
ATA1923/89 1989-08-11
AT192389 1989-08-11

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WO1991002421A1 true WO1991002421A1 (fr) 1991-02-21

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US (1) US5402449A (fr)
EP (1) EP0486554B1 (fr)
JP (1) JP3088454B2 (fr)
AT (1) ATE113432T1 (fr)
DE (1) DE59007578D1 (fr)
DK (1) DK0486554T3 (fr)
ES (1) ES2063372T3 (fr)
WO (1) WO1991002421A1 (fr)

Cited By (8)

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GR900100865A (el) * 1990-12-17 1992-11-23 Siemens Ag Μέ?οδος και συσκευή για μετατροπή ψηφιακα διαμορφωμένων σημάτων λήψεως από την περιοχή υψηλών συχνοτήτων.
EP0522998A3 (en) * 1991-07-09 1993-07-28 Ascom Tech Ag Method for estimating the frequency offset in a quadrature receiver
EP0579100A1 (fr) * 1992-07-14 1994-01-19 Daimler-Benz Aerospace Aktiengesellschaft Procédé et dispositif de correction de phase en bande de base dans un récepteur "PSK"
EP0580243A3 (fr) * 1992-07-24 1994-02-09 Magnavox Electronic Systems Company Procédé et appareil d'excision d'une bande étroite de fréquence d'interférence pour la communication à spectre étalé
EP0597255A1 (fr) * 1992-11-07 1994-05-18 GRUNDIG E.M.V. Elektro-Mechanische Versuchsanstalt Max Grundig GmbH & Co. KG Récepteur pour radiodiffusion numérique avec traitement numérique de signaux
WO1994028662A1 (fr) * 1993-06-02 1994-12-08 Nokia Telecommunications Oy Procede de demodulation d'un signal module numeriquement et demodulateur
WO1997013349A3 (fr) * 1995-09-29 1997-06-12 Siemens Ag Configuration d'un recepteur destine a recevoir des signaux porteurs a modulation/manipulation angulaires de differentes frequences
US6510313B1 (en) 1997-11-07 2003-01-21 Koninklijke Philips Electronics N.V. Wireless communication device

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US5757857A (en) * 1994-07-21 1998-05-26 The Regents Of The University Of California High speed self-adjusting clock recovery circuit with frequency detection
US5832043A (en) * 1995-04-03 1998-11-03 Motorola, Inc. System and method for maintaining continuous phase during up/down conversion of near-zero hertz intermediate frequencies
US5687163A (en) * 1995-06-07 1997-11-11 Cirrus Logic, Inc. Method and apparatus for signal classification using I/Q quadrant histogram
US5661485A (en) * 1995-09-08 1997-08-26 Condor Systems, Inc. Homodyne receiver apparatus and method
US5802112A (en) * 1996-01-16 1998-09-01 Transcendat Inc. Multi-level, multi-frequency interference pattern analog waveform encoding of digital data for transmission
US6026129A (en) * 1996-03-27 2000-02-15 Matsushita Electric Industrial Co., Ltd. Radio receiving apparatus for receiving communication signals of different bandwidths
US6359944B1 (en) * 1996-04-17 2002-03-19 Thomson Licensing S.A. Tuning system for achieving quick acquisition times for a digital satellite receiver
US5937341A (en) 1996-09-13 1999-08-10 University Of Washington Simplified high frequency tuner and tuning method
DE19733732C2 (de) * 1997-08-04 1999-05-12 Siemens Ag Verfahren zur Unterstützung der einfachen Synchronisierung auf den Träger eines energieverwischten QPSK-Signals
US6038262A (en) * 1998-06-09 2000-03-14 Transcendata, Inc. Method and apparatus for compensation of electro-magnetic distortion using weighted feedback delay for amplitude coded sinusoidal waveform generation and transmission
DE19942944A1 (de) * 1999-09-08 2001-03-22 Infineon Technologies Ag Kommunikationssystem und entsprechender Empfänger
JP3576880B2 (ja) * 1999-09-09 2004-10-13 日本電気株式会社 自動変調方式識別装置及び自動変調方式識別方法
US7088765B1 (en) * 2000-03-15 2006-08-08 Ndsu Research Foundation Vector calibration system
US20010055348A1 (en) * 2000-03-31 2001-12-27 Anderson Christopher L. Sequential quadrant demodulation of digitally modulated radio signals
DE10120702A1 (de) * 2001-04-27 2002-11-14 Infineon Technologies Ag Hochfrequenzempfänger
US7272375B2 (en) * 2004-06-30 2007-09-18 Silicon Laboratories Inc. Integrated low-IF terrestrial audio broadcast receiver and associated method
US7333051B2 (en) * 2004-11-19 2008-02-19 Lockheed Martin Corporation Methods and devices for determining the linearity of signals
US8983003B2 (en) 2010-03-31 2015-03-17 Hytera Communications Corp., Ltd. Method and system for adaptively identifying signal bandwidth
CN112398768B (zh) * 2019-08-19 2024-01-16 博通集成电路（上海）股份有限公司 用于校准频偏的接收机和方法

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Cited By (10)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
GR900100865A (el) * 1990-12-17 1992-11-23 Siemens Ag Μέ?οδος και συσκευή για μετατροπή ψηφιακα διαμορφωμένων σημάτων λήψεως από την περιοχή υψηλών συχνοτήτων.
EP0522998A3 (en) * 1991-07-09 1993-07-28 Ascom Tech Ag Method for estimating the frequency offset in a quadrature receiver
EP0579100A1 (fr) * 1992-07-14 1994-01-19 Daimler-Benz Aerospace Aktiengesellschaft Procédé et dispositif de correction de phase en bande de base dans un récepteur "PSK"
EP0580243A3 (fr) * 1992-07-24 1994-02-09 Magnavox Electronic Systems Company Procédé et appareil d'excision d'une bande étroite de fréquence d'interférence pour la communication à spectre étalé
EP0597255A1 (fr) * 1992-11-07 1994-05-18 GRUNDIG E.M.V. Elektro-Mechanische Versuchsanstalt Max Grundig GmbH & Co. KG Récepteur pour radiodiffusion numérique avec traitement numérique de signaux
WO1994028662A1 (fr) * 1993-06-02 1994-12-08 Nokia Telecommunications Oy Procede de demodulation d'un signal module numeriquement et demodulateur
US5598125A (en) * 1993-06-02 1997-01-28 Nokia Telecommunications Oy Method for demodulating a digitally modulated signal and a demodulator
JP3403198B2 (ja) 1993-06-02 2003-05-06 ノキア テレコミュニカシオンス オサケ ユキチュア デジタル変調された信号を復調する方法及び復調器
WO1997013349A3 (fr) * 1995-09-29 1997-06-12 Siemens Ag Configuration d'un recepteur destine a recevoir des signaux porteurs a modulation/manipulation angulaires de differentes frequences
US6510313B1 (en) 1997-11-07 2003-01-21 Koninklijke Philips Electronics N.V. Wireless communication device

Also Published As

Publication number Publication date
EP0486554B1 (fr) 1994-10-26
ATE113432T1 (de) 1994-11-15
ES2063372T3 (es) 1995-01-01
DK0486554T3 (da) 1994-12-27
JPH05500888A (ja) 1993-02-18
JP3088454B2 (ja) 2000-09-18
US5402449A (en) 1995-03-28
DE59007578D1 (de) 1994-12-01
EP0486554A1 (fr) 1992-05-27

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