JPH10242932A - Diversity receiver for frequency-division multiplex signal - Google Patents

Abstract

PROBLEM TO BE SOLVED: To stably execute the diversity reception of a frequency-division multiplex signal under a multi-pass environment. SOLUTION: A matrix circuit 15 generate a plurality of signals having mutually different synthetic ratios expressed by mA+(1-m)B} (provided that 0<=m<=1) from high frequency OFDM signals individually received by antennas 11, 12 arranged at an interval more than the half wavelength of a received radio wave and outputs these generated signals parallel to a maximum level selector 16. The selector 16 selects a signal of a maximum level from those plural input signals. Consequently discontinuity can be reduced, the synchronizing state of a receiver and the corrective follow-up of a transmission characteristic can be smoothly executed and continuous reception can be attained.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a diversity receiver for frequency division multiplexed signals, and more particularly to an orthogonal frequency division multiplexed signal (OFDM) based on multi-level QAM.
The present invention relates to a diversity receiver that receives uency division multiplexes using two antennas.

[0002]

2. Description of the Related Art In a device for receiving a signal wirelessly transmitted using radio waves in the VHF band or the like, not only a direct wave received by propagating directly from a transmitting side through a transmission line but also a reflection in the transmission line. An indirect wave reflected by the object is generated, which interferes with the direct wave to generate a multipath, and the transmission characteristics fluctuate. The transmission characteristics under this multipath environment fluctuate greatly especially in mobile transmission and reception. To reduce that effect,
2. Description of the Related Art Conventionally, there has been known diversity reception in which switching is performed to select an antenna having a good reception state from among a plurality of antennas installed spatially at a distance equal to or longer than a half wavelength of a transmission frequency. In normal diversity reception, reception is switched so as to select an antenna having the highest signal strength from among a plurality of antennas.

Conventionally, signals from a plurality of antennas are received by a plurality of receivers, an error rate of a signal in each reception is detected, and an antenna switch is switched so that the error rate is minimized. There is also known a diversity receiving apparatus that receives while selecting (Japanese Patent Laid-Open No. 5-29992,
JP-A-5-29993). Also known is a diversity receiver that uses a plurality of devices consisting of an antenna and a receiver and selects a group of these devices while switching to a group of devices that gives the maximum average received power or the minimum waveform distortion. Japanese Unexamined Patent Publication No. Hei 8-65222: Title of the invention: "Same frequency channel time division bidirectional transmission system").

Furthermore, in a communication system using a plurality of carrier frequencies, a plurality of frequency converters are used to share the center frequencies of different carriers in a phase comparison circuit, and the frequency-converted received signals are added at a fixed ratio. A diversity receiving apparatus that combines and reduces signal distortion due to multipath by averaging processing is also conventionally known (Japanese Patent Laid-Open No. 5-183540: title of invention "in-phase combined space diversity receiving apparatus"). At this time, a band filter for removing adjacent carrier components, an automatic gain control amplifier, and a voltage comparator are devised so as to be commonly used for received signals having different frequencies.

On the other hand, there is an OFDM signal transmission method as a method for transmitting an information signal by dividing it into a plurality of carriers.
In the OFDM signal transmission method, since information signals are transmitted using a large number of carriers, the speed of information transmitted on each carrier can be reduced, and the efficiency of the transmission frequency band can be increased by making the transmission spectrum in the frequency domain rectangular. This is a possible transmission method.

[0006] Usually, the carriers constituting the OFDM signal are modulated by the quadrature phase modulation (QPSK) system. In order to further increase the band utilization rate, a method of modulating each carrier by the multi-level quadrature amplitude modulation (QAM) is used. Is also taken. However, when performing communication by converting a modulated signal into multi-valued data, it is necessary to maintain a high-quality transmission path, and a device for that purpose is required.

[0007]

SUMMARY OF THE INVENTION A diversity reception system can be used for improving a transmission system accompanied by transmission distortion due to multipath. However, the diversity reception method of switching to the reception signal of the antenna with the best reception state among a plurality of antennas and receiving the same is directly applied to the reception of an OFDM signal in which a number of carriers modulated by multilevel QAM are frequency-division multiplexed. In this case, a change in the signal level at the time of switching the antenna becomes a problem.

As an example, OFDM using 256QAM
The stability of the signal required for the QAM demodulator in the signal transmission method is considered as follows. That is, transmission of information by each carrier is defined and performed within a 16 × 16 signal point arrangement plane. For this reason, the information point arranged in the center of the unit of each 16 × 16 signal point arrangement is arranged at the boundary with the adjacent signal point due to an error of 1/32. Therefore, it is necessary that the characteristics (stability) of the transmission path be further suppressed to a level fluctuation of about 1/64 which is a half thereof.

However, in practice, it is difficult to suppress the characteristics of the transmission path to a level fluctuation of about 1/64 as described above. Therefore, the applicant of the present invention has previously described Japanese Patent Laid-Open No. 7-336322
In Japanese Patent Laid-Open Publication No. H11-264, a method for performing demodulation of a multilevel QAM wave while performing calibration on each carrier (correcting transmission characteristics) has been disclosed. According to this method, the QAM is used together with the calibration carrier for transmitting the information signal.
A reference circuit for decoding is transmitted, and in a decoding circuit in the receiving device, the signal level of each real part and imaginary part of each frequency obtained by performing a fast Fourier transform (FFT) operation on the received signal, and the demodulation output of the reference carrier , The level of the quantized digital signal transmitted by the digital information transmission calibration carrier is obtained, and the digital information is decoded. According to the receiver proposed by the present applicant, transmission characteristics can be corrected at the time of mobile reception at a low speed.

At the time of mobile reception, in addition to an incoming direct wave, an indirect wave that is reflected and arrives at various objects in a transmission path, that is, a multipath signal component that changes with time is added, and an incoming radio wave is received. Is received as a composite wave of a plurality of signals having a time difference passing through the plurality of paths, that is, varying in phase amount. For example, a signal transmitted by 256QAM requires a carrier-to-noise power ratio (C / N ratio) 18 dB better than that of transmission by QPSK to decode the signal.

However, when the phase relationship between the direct wave and the multipath signal is in phase, it is good, but when the phase relationship is out of phase, they cancel each other out, and the signal strength of the arriving radio wave becomes very weak. , The required C / N ratio cannot be obtained, and the demodulated output of the reference carrier also decreases.
Due to the noise voltage (thermal noise) occurring in the receiver in the natural world, a good S / N ratio for the received signal cannot be secured, making reception difficult.

In this case, a method of correcting a signal that cannot be decoded by an error compensating circuit can be considered. However, since the time when the direct wave is canceled by the multipath signal and the signal level is lowered is relatively long, the signal cannot be transmitted during that time. In order to compensate the signal by the error compensation circuit, the signal to be transmitted needs to have a long interleave length. However, if the interleave length is increased in order to correct a signal that cannot be transmitted by the error compensating circuit, the delay time from transmission to reception increases, which makes it practically difficult to use the error compensating circuit.

The present invention has been made in view of the above points,
An object of the present invention is to provide a frequency division multiplexed signal diversity receiving apparatus capable of stably receiving a frequency division multiplexed signal under a multipath environment.

It is another object of the present invention to provide a frequency division multiplexed signal diversity receiving apparatus capable of optimally receiving an orthogonal frequency division multiplexed signal while reducing the influence on a calibration circuit.

[0015]

SUMMARY OF THE INVENTION To achieve the above object, the present invention provides first and second antennas for receiving orthogonal frequency division multiplexed signals, respectively, and a second antenna obtained by receiving the first antenna. 1 and the second received signal obtained by receiving the signal through the second antenna, outputting a plurality of combined signals having predetermined combining ratios different from each other, and combining the first and second received signals.
Synthesizing means for outputting a received signal of, a selecting means for selecting a signal of the maximum level from the output signals of the synthesizing means, and a demodulating means for orthogonally demodulating the signal selected by the selecting means,
And decoding means for decoding the digital information signal by discretely Fourier-transforming the output signal of the demodulation means.

According to the present invention, not only the first and second received signals obtained from the first and second antennas but also a plurality of synthesized signals obtained by synthesizing them at different synthesizing ratios from each other. Since a signal having a large signal level is selected and output to the demodulation means, discontinuity is small, the synchronization state of the receiving device, the correction of the transmission characteristics can be smoothly followed, and continuous reception can be performed.

According to another aspect of the present invention, there is provided a first and second antennas for receiving orthogonal frequency division multiplexed signals, respectively, and a first reception antenna obtained by receiving the signals on the first antenna. Signal and a second antenna received by the second antenna.
A plurality of synthesized signals having different predetermined synthesizing ratios generated by synthesizing the received signals, a synthesizing / selecting means for selecting a signal having a maximum level from the first and second received signals, A demodulation means for orthogonally demodulating the signal selected by the means, and a decoding means for discretely Fourier-transforming the output signal of the demodulation means to decode the digital information signal.

In the present invention, the combining / selecting means generates a plurality of combined signals obtained by combining the first and second received signals obtained by the first and second antennas at different combining ratios. And they and the first and second
Since the signal having the highest signal level is selected from the received signals and output to the demodulation means, the discontinuity is small, the synchronization state of the receiving apparatus, the correction of the transmission characteristics can be smoothly performed, and the continuous Can receive.

Further, the first and second antennas according to the present invention are characterized in that they are spaced apart from each other by at least 1/2 times the wavelength of the center carrier of the orthogonal frequency division multiplexed signal in the high frequency band to be received. I do. Accordingly, in a multipath environment, the phase relationship between the direct wave and the indirect wave in the first and second antennas is different, and even if the reception intensity of one antenna is greatly attenuated, the direct wave and the indirect wave are in the other antenna. Since the wave phase relationship is different, an appropriate signal strength is often obtained.

[0020]

Next, embodiments of the present invention will be described with reference to the drawings. FIG. 1 is a block diagram showing an embodiment of a frequency division multiplexed signal diversity receiving apparatus according to the present invention. This embodiment is a receiving apparatus for receiving an OFDM signal transmitted by an orthogonal frequency division multiplexing signal transmission method proposed by the present applicant in Japanese Patent Application No. 8-43854. That is, in the orthogonal frequency division multiplexing signal transmission scheme proposed by the present applicant, a reference signal is inserted into an OFDM signal as known reference data on the transmission side and transmitted, and the reception side performs A transmission path characteristic indicating an error between an I signal and a Q signal is detected, and a correction equation is obtained to correct the detected transmission path characteristic.

Here, of a plurality of carriers constituting the OFDM signal, one positive carrier and one negative carrier which are symmetrical to each other and which are affected by crosstalk with respect to the center carrier. , And the reference signal is transmitted on the set of these carriers. Further, here, the positive / negative carrier for inserting the reference signal is designated by a symbol number, and the set of positive / negative carriers is switched at regular time intervals to detect the transmission path characteristics of all carriers. is there.

Here, OFD transmitted by the transmitting device
To explain the outline of the M signal, the transmitting apparatus separately modulates an information signal to be transmitted as a signal including a real part and an imaginary part, and modulates an in-phase signal (I signal) and a quadrature signal (Q signal). Generates and modulates a plurality of positive carriers on the high frequency side and a plurality of negative carriers on the low frequency side with respect to the center carrier, among a plurality of carriers having different frequencies in the in-phase signal and the quadrature signal, and , Frequency division multiplexed,
A multi-level QAM OFDM signal is generated and transmitted. A known reference signal is transmitted on a predetermined carrier among the plurality of carriers, and the predetermined carrier is changed cyclically at regular intervals. To send.

Specifically, for example, a symbol number is inserted into a specific carrier, and a known reference signal (reference data) is inserted into another positive or negative carrier corresponding to the symbol number. The above symbol numbers are 0, 1, 2,
3,. . . , 511, 0, 1, 2,. . And so on
This is a 9-bit symbol number that sequentially and cyclically changes every symbol period. Since accurate decoding of this symbol number is important, dedicated reference data (reference signal) is prepared, and a multi-level QAM (256 QA) used for another carrier is used.
M), multi-level modulation with a smaller number of multi-levels is performed. Specifically, of the above 9 bits representing the symbol number,
The four bits of the ninth, eighth, third and second bits are transmitted and received by 16QAM. On the receiving side, the symbol number from these 4 bits is changed to 9 bits. This decoding is easy because the symbol number is a sequence of numbers that are sequentially increased by one.

Also, of the 9-bit symbol number,
An inverse discrete Fourier transform (IDFT) operation is performed so that the fourth bit of the ninth, eighth, third, and second bits is transmitted on a specific carrier (for example, the first carrier), and is obtained based on the symbol number of the upper seven bits. An IDFT operation is performed so that the reference signal is transmitted on another specific carrier. Since the lower two bits of the symbol number are ignored, a reference signal having the same value is inserted between four symbols.

On the basis of the least significant bit of the 9-bit symbol number, the symbols are divided into odd symbols and even symbols.
The following two types of reference signals are inserted.

[0026]

(Equation 1) The determinant reference signal represented by the equation (1a) is inserted into an even symbol, and the determinant reference signal represented by the equation (1b) is inserted into an odd symbol. Here, X is a known reference signal value. The carrier frequency for inserting the reference signal is transmitted as a set of positive and negative carrier frequencies symmetric with respect to the center carrier frequency F0.

Next, the embodiment of the present invention shown in FIG. 1 will be described. The above-mentioned OFDM signal transmitted as a signal in the high frequency band from the transmitting device is transmitted through the transmission line to the antenna 11.
And 12. Here, antennas 11 and 1
Reference numeral 2 is provided at a position separated from each other by a half wavelength or more of the center carrier (for example, one hundred and several tens of MHz) of the above-mentioned OFDM signal in the high frequency band.

This is for the following reason. In a reception environment where many multipath signals exist, reception antennas 11 and 1
The distribution of the electric field strength near 2 greatly fluctuates. At positions that differ by more than half wavelength from the carrier to be transmitted, the phase relationship between the direct wave and the indirect wave is also different. Even if the reception strength of one antenna is greatly attenuated, an appropriate signal strength is obtained in the other antenna. Often. That is, the antennas 11 and 12 which are arranged at a distance of at least half a wavelength from the carrier frequency spatially are different from each other even when one of the antennas receives the direct wave and the indirect wave by canceling each other. This is because, at the position, the phase relationship between the direct wave and the indirect wave is different.

In other words, even when the direct wave and the indirect wave cancel each other out in the antenna 11 (or 12) and the received signal strength becomes very weak, the direct wave in the antenna 12 (or 11) installed at a position different by half a wavelength or more is used. The phase relationship between the wave and the indirect wave is often different from the relationship in the antenna 11 (or 12), and it can be expected that the phase relationship between the direct wave and the indirect wave does not cancel each other. Therefore, antenna 1
By connecting different OFDM signal demodulators to each of 1 and 12, and selecting and obtaining data with less error than the demodulated outputs of both, a received output with minimal influence of multipath can be obtained.

However, the multi-level QAM OFDM signal receiving apparatus has another problem different from the above. That is,
In a multi-level QAM OFDM signal receiving apparatus, a synchronization state is established after the OFDM signal is input, and it takes a considerable time (for example, several seconds) for adaptively operating a transmission path characteristic compensation circuit while decoding. ). At the receiving point, when the multipath signal is fluctuating, the signal level becomes equal to or lower than a predetermined value, and when the receiving device is out of synchronization, reception becomes impossible. A considerably long time is required, and the receiving input signal cannot be decoded during this long time until the receiving apparatus operates in a synchronized state and the adaptive compensation circuit for the transmission line characteristic operates normally.

That is, after performing the FFT operation on the OFDM signal, decoding of the signal point arrangement is performed using the calibrated table. However, when the signal is switched, the level of the input signal, the rotation of the IQ axis, etc. Until a calibration table corresponding to a new signal is created, the signal cannot be decoded, and time is required for that.

As described above, in the diversity reception of the multi-level QAM OFDM signal in the receiving apparatus, it is necessary to continuously secure the synchronization state of the receiving apparatus and the correction follow-up of the transmission path characteristics without interrupting. . Therefore, in this embodiment, the high-frequency OFDM signals separately received by the antennas 11 and 12 are respectively combined with the antenna 1
After being supplied to and amplified by input amplifiers 13 and 14 provided corresponding to 1 and 12, the signal is supplied to a matrix circuit 15.

The matrix circuit 15 receives the first reception signal A received by the antenna 11 and amplified and extracted by the input amplifier 13, and the second reception signal A received by the antenna 12 and amplified and extracted by the input amplifier 14. After the signal B is divided, the signal B is combined and output at a predetermined combination ratio. That is, the matrix circuit 15
A + (1−m) B｝ (where 0 ≦ m ≦ 1), and generates a plurality of signals having different combining ratios, and outputs these signals to the maximum level selector 16 in parallel.

The maximum level selector 16 selects the signal of the maximum level from the plurality of input signals. At this time, for example, when there are a plurality of signals whose level is higher than the currently selected signal, the signal of the gradually higher level is switched and selected from the currently selected signal, and finally, the signal of the highest level is finally selected. The signal may be switched and selected.

Here, the maximum level selector 16 is
A synchronizing signal generating circuit 22 to be described later is performed such that an FFT and QAM decoding circuit 26 to be described later is performed at the beginning of a guard interval period in which an FFT operation is not performed among symbol periods including a guard interval period and an effective symbol period.
The above-mentioned switching is performed based on the symbol synchronization signal from.

The received signal selected by the maximum level selector 16 is decoded in the same manner as the orthogonal frequency division multiplexed signal receiving apparatus disclosed in Japanese Patent Application No. 8-43854 previously proposed by the present applicant. You. That is, the received signal selected by the maximum level selector 16 is
Is converted to an intermediate frequency by the intermediate frequency amplifier 1
After being amplified by 8, it is supplied to a carrier extraction and quadrature demodulator 19 having a configuration described later.

The carrier extraction circuit portion of the carrier extraction and quadrature demodulator 19 is a circuit for extracting the center carrier (hereinafter, also referred to as a carrier) of the input OFDM signal as accurately as possible with a small phase error. Here, each carrier for transmitting information is arranged adjacent to every 387 Hz, which is a symbol frequency, to form an OFDM signal. Therefore, the carrier for information transmission adjacent to the center carrier is also 3% away from the center frequency.
87 Hz apart, and to extract the center carrier,
A circuit having a high degree of selectivity is required so as not to be affected by adjacent carriers for information transmission which are separated only by 387 Hz.

The center carrier F0 extracted by the carrier extraction and quadrature demodulator 19 is supplied to an intermediate frequency oscillator 20, where it is 10.7 phase-locked to the center carrier F0.
Generate an intermediate frequency of MHz. Intermediate frequency oscillator 2
The output intermediate frequency of 0 is directly supplied to the quadrature demodulator 19 as the first demodulation carrier, and the carrier is extracted and quadrature demodulated as the second demodulation carrier after the phase is shifted by 90 ° by the 90 ° shifter 21. Is supplied to the vessel 19.

Thus, an analog signal (frequency division multiplexed signal) equivalent to the analog signal input to the quadrature modulator of the transmitting device is demodulated and extracted from the carrier extraction and quadrature demodulator unit of the quadrature demodulator 19. While being supplied to the synchronizing signal generation circuit 22, a signal of a required frequency band transmitted as OFDM signal information by a low-pass filter (LPF) 23 is passed, supplied to an A / D converter 24, and converted into a digital signal. Is done.

What is important here is the sampling timing for the input signal of the A / D converter 24, which is based on a sample synchronization signal having a frequency twice the Nyquist frequency, which is generated from the pilot signal by the synchronization signal generation circuit 22. Generated. That is, the pilot signal is set at a predetermined integer ratio with respect to the sample clock frequency, and the frequency of the pilot signal is multiplied according to the frequency ratio to obtain the timing of the sample clock.

The synchronizing signal generating circuit 22 receives a demodulated analog signal, and generates a sample synchronizing signal by a PLL circuit which performs phase synchronization with a pilot signal transmitted as a continuous signal in each symbol period including a guard interval period. A generating circuit section, a symbol synchronization signal generating circuit section for examining a phase state of a pilot signal based on a signal extracted from a part of the sample synchronization signal generation circuit section, detecting a symbol period, and generating a symbol synchronization signal; A system clock generating circuit for generating a system clock such as a section signal for removing a guard interval period from the signal and the symbol synchronization signal.

The digital signal extracted from the A / D converter 24 is supplied to the guard interval period processing circuit 25, where the digital signal is less affected by multipath distortion based on the system clock from the synchronization signal generation circuit 22. Is obtained and supplied to the FFT / QAM decoding circuit 26.

The FFT (Fast Fourier Transform) circuit section of the FFT / QAM decoding circuit 26 performs a complex Fourier operation using the system clock from the synchronizing signal generation circuit 22 and outputs the output signal of the guard interval period processing circuit 25 for each frequency. The signal levels of the real part and the imaginary part are calculated.

The real part for each frequency obtained by the above,
Each signal level of the imaginary part is compared with the demodulated output of the reference carrier by the QAM decoding circuit section, whereby the level of the quantized digital signal transmitted by the carrier for digital information transmission is obtained. Decrypted. The decoded digital information signal is subjected to output processing such as parallel-serial conversion by the output circuit 27 and is output to the output terminal 28.

Here, the FFT / QAM decoding circuit 26
For example, the configuration is as shown in the block diagram of FIG.
In the figure, the I signal I ′ and the Q signal Q ′ input from the guard interval period processing circuit 25 are supplied to an FFT symbol number decoding circuit 261.

[0046] By the way, a complex number that is assigned to a carrier frequency _{+ W n (p + jq)} , center the carrier F0
The complex number (r + ju) assigned to the negative carrier frequency −W _n symmetrical to the carrier frequency + W _n is subjected to IDFT calculation on the transmission side to produce a time-axis waveform I signal and Q signal, which are In addition to the influence of the OFDM signal transmission system including the influence in the multipath environment, the signal undergoes amplitude change and phase change, and is demodulated into the above-mentioned demodulated signal I 'signal and Q' signal by the quadrature demodulator of the receiver. Is done. The demodulated signal I 'is represented by (p' + jq '), and the demodulated signal Q' is represented by (r '+ ju'). They are,
When the coefficients of S0 to S7 are newly introduced and organized, the I
The signal and Q signal have the following relationship.

(P ′ + jq ′) = (p + jq) (S0 + jS1) + (r−ju) (S2 + jS3) (2) (r ′ + ju ′) = (r + ju) (S6 + jS7) + (p−jq) (S4 + jS5) (3) That is, when Expressions (2) and (3) are expressed using a determinant,

[0048]

(Equation 2) It is.

From this, S0 indicates the rate at which the real part of the positive carrier is transmitted to the real part of the positive carrier, and indicates the rate at which the imaginary part of the positive carrier is transmitted to the imaginary part of the positive carrier. S1 indicates the rate at which the real part of the positive carrier leaks to the imaginary part of the positive carrier, and the rate at which the imaginary part of the positive carrier leaks to the real part of the positive carrier. S2 indicates the rate at which the real part of the negative carrier leaks to the real part of the positive carrier, and the imaginary part of the negative carrier is
The ratio of leakage to the imaginary part of positive carriers is shown.

S3 indicates the rate at which the real part of the negative carrier leaks to the imaginary part of the positive carrier, and the rate at which the imaginary part of the negative carrier leaks to the real part of the positive carrier. S4 indicates the rate at which the real part of the positive carrier leaks to the real part of the negative carrier, and the imaginary part of the positive carrier is
The rate of leakage of negative carriers to the imaginary part is shown. S5
Indicates the rate at which the real part of the positive carrier leaks to the imaginary part of the negative carrier, and the rate at which the imaginary part of the positive carrier leaks to the real part of the negative carrier. S6 indicates the rate at which the real part of the negative carrier is transmitted to the real part of the negative carrier, and indicates the rate at which the imaginary part of the negative carrier is transmitted to the imaginary part of the negative carrier. S7 indicates a rate at which the real part of the negative carrier leaks to the imaginary part of the negative carrier,
This shows the rate at which the imaginary part of the negative carrier leaks to the real part of the negative carrier. Here, the description of-and + of the rate was omitted.

That is, the above-described coefficients S0 to S7 indicate the characteristics of the transmission path of the I signal and the Q signal, and by calculating these coefficients S0 to S7, the characteristics of the transmission path can be detected. Further, by obtaining the inverse matrix of equation (4), it is possible to correct the received data and estimate the transmitted data.

Therefore, in the embodiment of the present invention, as described above, a symbol number is inserted into a specific carrier, and a known reference signal (reference data) is inserted into another positive or negative carrier corresponding to the symbol number. I do. The carrier frequency at which the reference signal is inserted and transmitted is determined in advance in association with the symbol number, and is switched at regular intervals. This is because transmission characteristics are often different for each frequency.

The I signal I 'and Q signal Q' are F
After the FT symbol number decoding circuit 261 decodes the symbol number, a reception reference signal value of a set of positive and negative carriers corresponding to the symbol number is obtained next.

In this embodiment, since the predetermined 4 bits of the symbol number are subjected to 16QAM, the symbol number can be decoded with a better error rate than other transmission information, and 0 to 9 bits are represented. Symbol numbers up to 511 can be reliably decoded, and the insertion carrier of the transmission information reference signal can be reliably specified for each symbol.

In the 0th symbol (even symbol), the received reference signal value is p _0s ′ _, q _0s ′,
r _{0s ', u 0s',} in the first symbol (odd _{symbol), p 1s ', q 1s} ', r 1s ', u 1s'
In the second symbol (even symbol), p _2s ′
_, Q _2s ′, r _2s ′, u _2s ′, p _3s ′ _, q _3s ′,
r _3s ′ and u _3s ′.

The transmission line characteristic detection circuit 262 shown in FIG. 2 calculates the transmission line characteristic represented by the following expression by the 0th symbol and the first symbol based on the above expressions (4), (1a) and (1b). Are calculated.

S0 = ( _p0s ′ + _q1s ′) / (2X), S4 = ( _{r0s′− u1s} ′) / (2X) S1 = ( _{q0s′− p1s} ′) / (2X), S5 = ( _{u0s'- r1s} ') / (2X) S2 = ( _{p0s'- q1s} ') / (2X), S6 = ( _{r0s'- u1s} ') / (2X) S3 = (q _{0s '+ p 1s') /} (2X), S7 = (u 0s '-r 1s') / (2X) (5) 2 -th symbol and the third symbol is also in the same manner the coefficient S0~S7 determined by. Thereafter, the coefficients S0 to S7 are respectively averaged to remove white noise. In this embodiment, an average among four symbols is obtained.

Since there are eight coefficients S0 to S7, the coefficients can be obtained by transmitting and receiving two types of reference signals. Of course,
Since the reference signal is known in the receiving apparatus, any reference signal may be used as long as the coefficient can be obtained. In this way, the transmission path characteristic detection circuit 262 detects the transmission path characteristic coefficients S0 to S7 and supplies them to the first correction equation derivation and holding circuit 263 together with the symbol numbers.

The first correction formula deriving and holding circuit 263 first obtains the inverse matrix of the formula (4) from the inputted coefficients S0 to S7. This inverse matrix is expressed by the following equation.

[0060]

(Equation 3) Here, H0 to H7 and det A in the above equation are represented by the following equations.

H0 = + S0 (S6S6 + S7S7) -S2 (S4S6 + S5S7) + S3 (S4S7-S5S6) H1 = + S1 (S6S6 + S7S7) -S3 (S4S6 + S5S7) -S2 (S4S7-S5S6) H2 = + S4 (S2S2 + S3S3) -S6 (S0S2 + S1S3) + S7 (S0S3-S1S2) H3 = + S5 (S2S2 + S3S3) -S7 (S0S2 + S1S3) -S6 (S0S3-S1S2) H4 = + S2 ( S4S4 + S5S5) -S0 (S4S6 + S5S7) -S1 (S4S7-S5S6) H5 = + S3 (S4S4 + S5S5) -S1 (S4S6 + S5S7) + S0 (S4S7-S5S6) H6 = + S6 (S1S0 + S1) -S4 (S0S2 + S1S3) -S5 (S0S3-S1S2) H7 = + S7 (S0S0 + S1S1) -S5 (S0S2 + S1S3) + S4 (S0S3-S1S2) det A = S0 × H0 + S1 × H1 + S4 × H2 + S5 × H3 Subsequently, the first correction formula deriving and holding circuit 263 calculates det A and H0 to H7 based on the above formula, and further calculates the inverse matrix of the inverse matrix in formula (6) from these calculated values. The value of the following correction equation is calculated and stored.

[0062]

(Equation 4) In this way, a correction formula for the corresponding positive / negative carrier is prepared. The corresponding positive / negative carrier is determined by the symbol number. Naturally, there is a correction formula for each carrier, and when 257 carriers are used as in this embodiment, about 1
28 correction formulas are sequentially calculated and held. Since the first correction formula is obtained by averaging one corresponding positive / negative carrier with four symbols, the interval at which the next average correction formula of the same positive / negative carrier is updated is 512 symbols later (4
(Symbol x 128 sets = 512 symbols).

The first correction circuit 264 derives and holds the information received from the FFT symbol number decoding circuit 261 by the first correction formula derivation and holding circuit 263.
The following equation is calculated using the correction equation (1), and a corrected signal is output.

[0064]

(Equation 5) Where a ″ and b ″ are the real and imaginary parts of the corrected received data assigned to the positive carrier frequency, and c ″ and d ″ are assigned to the negative carrier frequency. Real and imaginary parts of the corrected received data, a ′
And b 'are the real and imaginary parts of the received data of the positive carrier frequency, and c' and d 'are the real and imaginary parts of the received data of the negative carrier frequency.

As described above, the received signal is corrected, and the corrected transmission information R ″ is output from the first correction circuit 264.
(A ", c") and I "(b", d "),
Second correction circuit 266 and second correction expression derivation holding circuit 2
65 respectively. Further, the symbol number is supplied from the first correction formula derivation and holding circuit 263 to the second correction formula derivation and holding circuit 265.

The second correction formula deriving and holding circuit 265
The corrected signal a ″ input from the correction circuit 264 of
b ″, c ″ and d ″ and the first correction formula deriving and holding circuit 26
3 and the second correction circuit 26
Based on the decoded data a, b, c, and d input from No. 6, a K matrix of the formula (11) is generated and
Is held as the correction equation. here,

[0067]

(Equation 6) Can be requested. Here, the left side of the equation (9) indicates that the transmission information signals a, b, c, and d have an error δ that is a fast changing component.
_p , δ _q , δ _r , δ _u , respectively, and correspond to information signals a ″, b ″, c ″, d ″ corrected by the first correction formula. Means In the equation (11), K0, K1, K6 and K7 are represented by the following equations.

K0 = (aa ″ + bb ″) / (a ″ ² + b ″ ² ) (12a) K1 = (ab ″ −a ″ b) / (a ″ ² + b ″ ² ) (12b) K6 = (cc ” + Dd ") / (c" ² + d " ² ) (12c) K7 = (cd" -c "d) / (c" ² + d " ² ) (12d) The above K matrix is generated for each reception carrier frequency, And,
Used for the following symbols:

The second correction circuit 266 uses the K matrix which is a unit matrix for the next symbol after receiving the reference signal, and the K matrix generated for the previous symbol for the other symbols, using the following equation.

[0070]

(Equation 7) The input signals a ″, b ″, c ″, and d ″ are further corrected (second correction) based on <a>, <b>, <c>, and <d
>, Based on which the decoded data a, b, c and d are generated and output to the output circuit 32 of FIG.

In this manner, the second correction circuit 266 corrects errors and characteristics that change relatively slowly, such as aging and temperature changes, by the first-stage correction using the first correction formula. In addition, accurate correction is performed by using a known reference signal, and subsequently, a relatively high speed such as a multipath environment generated in mobile communication or the like is corrected by a second-stage correction using a second correction formula. The decoded data a, which has been optimized for each symbol,
b, c and d (real part data R and imaginary part data I) are output.

In this embodiment, the maximum level selector 16 in FIG. 1 performs switching at the beginning of a guard interval period in which no FFT operation is performed, based on the symbol synchronization signal from the synchronization signal generating circuit 22. Therefore, the correction operation by the second correction circuit 266 can follow,
Diversity reception of an OFDM signal in a multipath environment can be suitably performed.

Next, another embodiment of the present invention will be described. In the above embodiment, the switching is performed so that the maximum received signal level is obtained by the maximum level selector 16. Normally, the received signal level does not change significantly during the symbol synchronization period. Depending on the situation, it may be greatly changed. In particular, in a receiving apparatus that receives a multilevel QAM OFDM signal in a multipath environment, as described above, even if a good signal is received, a considerably long time is required for the synchronization operation, so that the received signal level is significantly different. Then, each circuit of the receiving apparatus may not be able to follow it, in which case decoding cannot be performed. Therefore, in the present embodiment, the selection of the above-described maximum-level reception signal is switched so that the change in the reception signal level is small and smooth.

FIG. 3 is a block diagram showing a main part of another embodiment of the diversity receiving apparatus for frequency division multiplexed signals according to the present invention. In the figure, the variable level ratio mixer section 3
0 denotes signal dividers 31 and 32 and signal combiners 33, 3
4 and 35. Signal splitters 31 and 3
Reference numeral 2 denotes a configuration in which the division ratio is variably controlled by an external signal. The signal output to the output terminal 40 is shown in FIG.
, And then decoded in the same manner as described with reference to FIGS.

To explain the operation of this embodiment, the first received signal input from the antenna 11 of FIG. 3 via the input amplifier 13 is supplied to the signal divider 31, where it is divided into three signals. The signals are supplied to the synthesizers 33, 34 and 35, respectively. On the other hand, at the same time, the second received signal input from the antenna 12 via the input amplifier 14 is supplied to the signal divider 32, where it is divided into three and supplied to the signal combiners 33, 34 and 35, respectively. You.

Here, assuming that the level of the first received signal is A and the level of the second received signal is B, the signal divider 3132 calculates the level E1 obtained by performing the following operation.
1, E12 and E13 are output.

E11 = {(I + 1) / N} × A (14) E12 = (I / N) × A (15) E13 = {(I−1) / N} × A (16) Similarly, The signal divider 32 outputs signals of levels E21, E22, and E23 obtained by performing the following calculation.

E21 = {(NI-1) / N} × B (17) E22 = {(NI) / N} × B (18) E23 = {(NI−1) / N} × B (19) In the equations (14) to (19), N is a natural number that determines the minimum unit of the composition ratio, I is an integer set by the up / down counter 38, and is a value larger than -2N and smaller than + 2N. Take.

The signal combiner 33 adds and combines the signal of the level E11 from the signal divider 31 and the signal of the level E21 from the signal divider 32, and outputs a signal of the level F1. Similarly, the signal synthesizer 34 includes the signal divider 3
1 and the signal of the level E12
The signal of the level E22 is added and synthesized to output a signal of the level F2.
1 and the signal of the level E13
The signal of level F23 is added and synthesized with the signal of level E23 to output a signal of level F3.

Accordingly, the above signal levels F1, F2 and F3 are represented by the following equations.

F1 = [{(I + 1) / N} × A] + {(NI−1) / N} × B (20) F2 = {(I / N) × A} + {(NI) / N} × B (21) F3 = [{(I−1) / N} × A] + {(N−I + 1) / N} × B (22) It can be seen from equations (20) to (22). As described above, the signal level F1 has the largest combined ratio of the first received signal A to the second received signal B, and then decreases in the order of the signal levels F2 and F3.

The level comparator 36 shown in FIG.
3 and the level F3 output signal of the level comparator 35, and the count value of the up / down counter 38 is incremented or decremented by "1" based on the obtained level comparison result. I do. Here, the level comparison operation of the level comparator 36 is based on the symbol synchronization signal input via the terminal 37 and is based on the symbol transmission signal of the OFDM signal. Done in a period.

The count value I of the up / down counter 38
Is set to N / 2 in the initial state. Therefore, in this initial state, the signal level F2 output from the signal combiner 34 is obtained by substituting I = N / 2 into the equation (21), and F2 = (A + B) / 2. That is, in the initial state, a signal obtained by adding and averaging the first received signal A and the second received signal B is extracted from the signal combiner 34, and this is output via the terminal 40 as a signal compensator output signal through the terminal 40 in FIG. Is output to the frequency converter 17. In this case, under ideal transmission conditions free from the influence of multipath and the like,
This means that a signal having a good / N ratio is supplied to the demodulation circuit.

Assuming that there is a multipath component and the first received signal level A from the antenna 11 is reduced, the level F1 of the output signal of the signal combiner 33 is Compared to level F3 of (20)
As can be seen from the expressions and the expression (22), the value decreases. At this time, the level comparator 36 controls the up / down counter 38 so as to reduce the count value I of the up / down counter 38 by “1”.

The up / down counter 38 performs a counting operation (subtraction operation in this case) within the guard interval period according to a synchronization signal input via the terminal 39 every guard interval period, and counts the counter value I after the subtraction. It is supplied as a control signal to the signal dividers 31 and 32, respectively, and newly calculated signal levels E11 to E13, E2
1 to E23 are output.

Here, as can be seen from the equations (20) and (22), the second received signal of the first received signal level A at the signal level F1 is obtained by subtracting "1" from the counter value I. Since the mixing ratio of the first received signal level A at the signal level F3 to the second received signal level B at the signal level F3 decreases, the signal level F1 slightly increases from the immediately preceding value. , And the signal level F3 slightly decreases from the immediately preceding value.

The level comparator 36 compares the calculated signal levels F1 and F3, and if F1 <F3, the up / down counter 38 reduces the count value I of the up / down counter 38 again by "1". Control. As a result, the counter value I of the up / down counter 38 is further reduced by "1", the signals are again calculated by the signal dividers 31 and 32 based on the counter value after the subtraction, and further combined by the signal combiners 33 to 35. The output signal levels F1 to F3 are output.

Hereinafter, in the same manner as described above, the above operation is continued until the maximum output signal is detected, that is, in the level comparator 36, the signal levels F1 and F3 become equal. As described above, in this embodiment, a high-frequency reception signal whose maximum reception level constantly changes and gradually gradually changes to the maximum reception level is extracted from the signal combiner 34 to the output terminal 40.

In some transmission systems, the level of the multipath is higher than that of the direct wave. In such a case, when the received signals of the antennas 11 and 12 are added in opposite phases,
In some cases, a larger signal level is obtained. The circuit is operated with the value I of the up / down counter 38 exceeding the range of 0 to N because the antennas 11 and 12 include such good input signals having an antiphase relationship. This is because an input signal for a demodulator is obtained.

[0090]

As described above, according to the present invention,
A plurality of synthesized signals obtained by synthesizing the first and second received signals obtained by the first and second antennas at different synthesizing ratios respectively are generated, and a plurality of synthesized signals are generated from the first and second received signals. By selecting the signal with the highest signal level and outputting it to the demodulation means, the discontinuity is reduced, the synchronization status of the receiver and the correction of the transmission characteristics can be smoothly followed, and continuous reception is possible. Can receive a multi-level QAM orthogonal frequency division multiplexed signal more stably,
It is particularly suitable for application to mobile reception under a multipath environment.

Further, according to the present invention, the signal to be selectively output to the demodulation means is sequentially switched to the higher level to finally select the signal of the maximum level. The influence of the division multiplexed signal on the calibration circuit of the decoding means can be reduced, and the synchronization state of the receiver and the correction follow-up of the transmission characteristics can be continuously ensured without intermittent interruption.

Further, according to the present invention, the combined ratio of the first and second received signals can be applied not only to the positive-phase received signal but also to the inverted-phase received signal. Even when a phase component is strongly received, it is converted and added so as to have the same phase as the desired received signal, so that the demodulation unit can demodulate the signal while securing an appropriate input signal level.

[Brief description of the drawings]

FIG. 1 is a block diagram of an embodiment of the present invention.

FIG. 2 is a block diagram of an example of a main part in FIG.

FIG. 3 is a block diagram of a main part of another embodiment of the present invention.

[Explanation of symbols]

 Reference Signs List 11 first antenna 12 second antenna 13, 14 input amplifier 15 matrix circuit (synthesis means) 16 maximum level selector (selection means) 17 frequency converter 19 carrier extraction and quadrature demodulator (demodulation means) 22 synchronization signal generation Circuit (decoding means) 26 FFT, QAM decoding circuit (decoding means) 30 Variable level ratio mixer section (synthesis / selection means) 31 first signal splitter 32 second signal splitter 33 first signal synthesizer 34 Second signal synthesizer 35 Third signal synthesizer 36 Level comparator 38 Up / down counter

Claims (8)

[Claims] 1. A first and a second antenna for receiving an orthogonal frequency division multiplexed signal, a first received signal obtained by receiving on the first antenna, and a signal on the second antenna. Synthesizing means for synthesizing the second received signal obtained as described above, outputting a plurality of synthesized signals having predetermined synthesizing ratios different from each other, and outputting the first and second received signals; Selecting means for selecting a signal of the maximum level from the output signals; demodulating means for quadrature demodulating the signal selected by the selecting means; discrete Fourier transform of the output signal of the demodulating means to decode a digital information signal A frequency division multiplexed signal diversity receiving apparatus. 2. A symbol synchronization circuit for detecting a symbol period of the orthogonal frequency division multiplexed signal from an output demodulation signal of the demodulation means, wherein the selection means is configured to output the symbol period based on an output symbol synchronization signal of the symbol synchronization circuit. 2. The frequency-division multiplexed signal diversity receiving apparatus according to claim 1, wherein an operation of selecting and outputting the maximum level signal is performed in a guard interval period within a symbol period. 3. When there is one or more of the synthesized signals between the current signal level being selected and output and the detected maximum level, the selecting means sequentially outputs the synthesized signals to a large level. 2. The diversity receiving apparatus for frequency division multiplexed signals according to claim 1, wherein the signal is switched to the end and the signal of the maximum level is finally selected. 4. The antenna according to claim 1, wherein the first and second antennas are separated from each other by at least 倍 times the wavelength of a center carrier of the orthogonal frequency division multiplex signal in a high frequency band to be received. The diversity receiving apparatus for a frequency division multiplexed signal according to claim 1. 5. A first and second antenna for receiving an orthogonal frequency division multiplexed signal, a first received signal obtained by receiving on the first antenna, and receiving on the second antenna. A plurality of synthesized signals having different predetermined synthesizing ratios generated by synthesizing the obtained second received signal and a synthesizing unit for selecting a signal of the maximum level from the first and second received signals. Selecting means, demodulating means for orthogonally demodulating the signal selected by the combining / selecting means, and decoding means for performing a discrete Fourier transform on an output signal of the demodulating means to decode a digital information signal. Frequency division multiplexed signal diversity receiver. 6. The first signal dividing means for changing the level of the first received signal by an external control signal and outputting first to third synthesized signals having different levels from each other. A second signal divider for changing the level of the second received signal by an external control signal and outputting fourth to sixth combined signals having different levels from each other;
A first signal combiner for combining the first and fourth combined signals to output a seventh combined signal, and combining the second and fifth combined signals to output an eighth combined signal A second signal combiner for combining the third and sixth combined signals to form a ninth signal;
A third signal synthesizer that outputs a combined signal of the first and second signals, a level comparator that compares the levels of the seventh and ninth combined signals,
The count value is controlled based on the level comparison result of the level comparator, and the counter outputs the count value as the external control signal to the first and second signal dividers. [{(I + 1) / N} × A] +
{(NI-1) / N} × B (where N is a natural number that determines the minimum unit of the composition ratio, I is an integer set by the counter, and is a value larger than -2N and smaller than + 2N;
A is the level of the first received signal, B is the level of the second received signal: the same applies hereinafter), and the eighth synthesized signal is {(I / N) × A} + {(NI ) / N｝ × B, and the ninth composite signal is [{(I−1) / N｝ × A] ”.
+ {(N−I + 1) / N} × B, and the level comparator determines the value of the counter by a predetermined value when the level of the seventh combined signal is smaller than the level of the ninth combined signal. And the level of the seventh composite signal is reduced to the ninth level.
6. The frequency according to claim 5, wherein when the level is higher than the level of the synthesized signal, the value of the counter is increased by a predetermined value, and the eighth synthesized signal is output to the demodulation means as the selected signal. Diversity multiplex signal diversity receiver. 7. The level comparator performs a comparison operation within an effective symbol period during which the decoding unit performs a decoding operation in a symbol period of the orthogonal frequency division multiplexed signal, and the counter performs a guard interval in the symbol period. 7. The diversity receiving apparatus for frequency division multiplexed signals according to claim 6, wherein the counting operation is performed within a period. 8. The antenna according to claim 1, wherein the first and second antennas are separated from each other by at least 倍 the wavelength of a center carrier of the orthogonal frequency division multiplexed signal in a high frequency band to be received. The diversity receiving apparatus for a frequency division multiplexed signal according to claim 5.